EVBUM2124 - NCP1203GEVB A 70 W Low Standby Power Supply
Contents
1. Fswll 2 4 50 1 3 50 A n 7 2 50 i 1 50 i 500M H tH HJ Vramp 14 0 10 0 s H 6 00 3 EL 2 00 2 00 10 0U 30 0U 50 0U 70 0U 90 0U Figure 8 Simulations Show a Capacitor Voltage Ramping Up from a Few Hundred of mV Up to Nearly 5 V The numerical application gives a 484 uH inductance with a peak current of 2 36 A The NCP1200 incorporates a skip cycle feature that forces the controller to slice the switching pattern when the power supply drives light loads Depending on the system time constants the recurrence of the burst can enter the audible frequency range Since the default skip cycle takes place at one third of maximum peak current it is better to avoid working at high peak current in normal operation Should noise still appear in skip mode pinl lets you select a different lower skip level unfortunately to the detriment of the standby power generating less mechanical noise As a result we slightly increased the primary inductance to 700 uH to further limit the noise in standby operation 16 www BD fffcom ON NCP1203GEVB MOSFET Selection The MOSFET drain voltage sees in normal operation a maximum voltage of Lleak 18 VinDC max Vout V N Ip Clump The first term represents the maximum rectified DC voltage a2. 1 A 1 Connect the test setup as shown above 2 Apply an input voltage 85 Vac Vin 265 Vac Electronic load 19 V upto4 A Yokogawa power meter WT210 Multimeter 50 or 60 Hz 3 Connect electronic load on output connector Table 1 DESIRED RESULTS Measurements Conditions Reuts Comments Output Voltage Vin 100 Vac Vout 19 V 5 no load Input consumption at high Vin 230 Vac Pin 0 09 W 15 No load measurement can line and no load no load Vout 19 V fluctuate run WT210 in aver age mode Vin 230 Vac Vout 19 V 5 lload 3 6 A 4 Output voltage at low line Vin 85 Vac Vout 19 V 5 and full load lload 3 6 A www BD ff com ON Appendix A Bill of Material All resistors are 5 1 4 W SMD 1206 unless otherwise noted All SMD capacitors are 1206 SMD 16 V types unless otherwise noted All through hole electrolytic capacitors are radial types unless otherwise noted Manufacturer references are given for specific components only Table 2 BILL OF MATERIALS NCP1203GEVB www BD Tt tom ON Substi RoHS Desig Toler Foot Manufacturer Part tution Com nator Description Value ance print Manufacturer Number Allowed pliant R1 R7 Chip Resistor 18 kQ 0 25 W Vishay CRCW120618KJN Yes R2 Fusible Resistor 2 2 Q 5W Vishay CP00052R2JE14 Yes R1A B Power Leaded 39 kQ 3 0 W Vishay PR030203902G
3. Vbulk is the seat of a rising voltage equal to Vripple or 50 Vpp This corresponds to a brought charge Q of 4 Qbulk Vripple Cbulk 9mC From Figure 5 we can calculate the amount of charge Q drawn from the input by integrating the input current over the diode conduction time tc Qin ao o 5 The expression of igiode t is www BD ff com ON NCP1203GEVB tc t 6 Ipeak After proper integration it comes in 1 Qin 2 Ipeak tc If we now equate Qbulk and Qin and solve for Ipeak it comes Qbulk 2 7 Ipeak es or 6 A peak We can now evaluate the RMS current flowing through the diodes ic idiode t 2 dt 0 Ipeak 2 Fline 1 9 A VAC 90 Irms Fline We selected a KBU4J diode bridge 600 V 4 A for the rectifying function A small resistor or best an NTC can however be put in series to limit the surge current when you plug the SMPS in the AC outlet to less than the diode maximum peak current Ifsm or what the standard imposes you Thanks to these numbers we compute the apparent power at low line 1 9 A x 90 V 170 VA which compared to our 87 5 Watts of active power neglecting the input diode bridge and Cbulk losses gives a power factor of WX 9 PF VA OS conform to what we could expect from this kind of offline power supply Transformer Calculation Transformer calculation can be done in several manners a you evaluate ALL t
4. overload condition By detecting a current setpoint pushed to the maximum the internal logic takes the decision to enter into a safe burst operation auto recovering when the default leaves Precise overload levels can thus be implemented e Guaranteed operation at low output levels the Vcc delivered by an auxiliary winding moves with the power output level because a coupling exists between both windings When the supply is used in battery charging applications Vout can move depending on the charging state That is to say when the battery is nearly empty its voltage can be close to zero forcing Vout at ON Semiconductor hitp onsemi com EVAL BOARD USER S MANUAL this level Thanks to the natural secondary auxiliary reflection the primary auxiliary winding cannot maintain a sufficient voltage on the control IC Vcc collapses and puts the controller in trouble probably entering an hiccup mode similar to that of a startup sequence DSS being decoupled from Vout you never see that phenomenon As you can see the DSS offers interesting features but on the other hand it can sometimes compromise key design parameters Standby power and power dissipation are one of these e Standby power the DSS standby power contribution can easily be evaluated Vgy X Iayg with Iayg the current consumption taken by the controller and Vyy the high voltage supply rail If Iayg equals 1 mA then we have a standby power of 350 mW at a 3
5. 80 0 36 0 120 0 48 0 160 48 0 160 0 0 10 1k 10k 100 k 10 100 1k 10k 100 k Figure 12 Simulated Bode Plot of the Figure 13 Measured Open Loop Gain Current Mode Flyback with a Network Analyzer www BD Tt tom ON NCP1203GEVB ANSO LXE pepeeu jl adh LA dwog dwey dude nee ASL 8H Sco MI Io 0 zo 2 8 V H ZZZZNZ 8vLVNIL tO 09N9dO4 LW 8vIVNIL d 09 LYNN 8d Jamod SN dN 1401 99L 0 L xne SN dN JO SLO N Hu zv LL 1 Hil 002 d1 Jeuondo dAO A ooze an9 19 90 SO AN Occ 0229 c VIs un ol 9l y 0 S0 9 x AGL 11 B 00L02Hg8IN ta U M GA BE XZ 21 HW Ze X c WS80dg c Ld adh asny M S ou ecc indu resnun Figure 14 The Simulation Schematic for Our 70 W Current Mode Power Supply com ON mi co o T www DD NCP1203GEVB 100 120 140 160 180 200 220 240 Input Voltage VAC Figure 15 Line Regulation Is Excellent Thanks to Current Mode and a Good Open Loop DC Gain Board Final Results Standby Power Measured on an Infratek watt meter operated in Watt hour accumulation mode for best accuracy run length 30 minutes Vin 120 VAC Vout 16 76 V Iout 0 Pin 78 mW Vin 240 VAC Vout 16 76 V Iout 0 2 Pin 84 mW Line Regulation The array in Figure 15 shows the performance when the input voltage is moving between both range ends
6. As one can see current mode control with good open loop gain ensures a AVout less than 1 mV for a 212 VDC input variation 106 dB DC audio susceptibility Load Regulation By varying the load current between 11 W and 70 W it is possible to plot the load regulation of the board as shown in Figure 16 16 765 16 760 16 755 16 750 16 745 16 740 16 735 16 730 16 725 16 720 16 715 0 VAC 110 VAC T7 Output Voltage V 20 40 60 80 Output Power W Figure 16 Load Regulation at Two Different Input Voltages Efficiency We have designed two boards one using the auxiliary winding for best standby performance and another one with the Dynamic Self Supply DSS left normally working Because of the auxiliary winding it has been necessary to further clamp the drain voltage in order to improve the primary overload detection It is not necessary with the DSS and therefore the RCD drain clamp network can be less aggressive thus slightly improving the efficiency Board 2 also features a 6 A MOSFET compared to a 3 A MOSFET on board 1 Board 1 aux winding Vin 110 VAC n 79 Vin 240 VAC n 83 5 Vin 110 VAC 1 83 4 Vin 240 VAC n 84 8 Board 2 DSS www BD Tt tom ON NCP1203GEVB TEST PROCEDURE Vout lout Electronic load WN e MZ 2 TINS Figure 17 Test Setup Required Equipment Test Procedure ac power supply 85 265 Vac
7. BVdss and the diode maximum reverse voltage A Schottky diode represents a good choice especially with a power supply that can possibly enter Continuous Conduction Mode CCM The lack of reverse recovery loss and a low forward drop play in favor of this component However because of the metal silicon junction moderate breakdown voltages are available for a moderate cost The MBR20100 represents an interesting choice since it welcomes two 100 V Schottky in a TO 220 package Being in thermal contact a parallel wiring is possible The 100 V VgnyM lets us calculate the minimum turn ratio we can go down to keeping an acceptable safety margin _ PIV Vout 12 VinDC max Np Ns lt 1 0 221 A final ratio of 1 0 166 offers an adequate safety margin Vreverse 80 V max The diode s conduction power is evaluating using the following formula 13 Pdiodeayg Vf Idavg Rd Idrms Rather than manually calculating these numbers we will see later on how a Spice simulator can do the job for us Primary Inductance and Peak Current For AC DC adapters delivering this amount of power in a small place it is of common practice to make the power supply enter CCM in the middle of the total AC range around 180 VAC in our case When the input AC voltage diminishes the on time increases and the primary secondary RMS current go up This implies a greater heatsink for the MOSFET but also larger aluminum cans for the secondary filters F
8. Let s us adopt a 40 ripple level or a 50 V drop from the corresponding Vinpeak To evaluate the equivalent load current which discharges Cbulk between the peaks we divide the input power by the average rectified voltage Pin Pout 1 lload Vrectavg n Vpeak weer 860 mA DC 90 VAC input voltage Thanks to Figure 4 information we can evaluate the capacitor value which allows the drop from Vpeak down to Vavg Vripple 2 to stay within our 50 V target dV C iload dt Pout e Vripple Cbulk 2 n Fline Vripple Vpeak Eur 171 uF or 180 uF for a normalized value Fline 50 Hz worse case Diode Bridge Selection To select the right rectifiers it is necessary to know the RMS current flowing through its internal diodes Prior to reach this final result we need to evaluate the diode conduction time From Figure 5 we can see that the diode starts to conduct when Vacin reaches Vmin and stops when reaching Vinpeak Idiode X 2 ms div Figure 5 When Vacin Reaches Vpeak the Diode Stops Conducting From a mathematical point of view we can calculate the time VAcin takes to reach Vmin with Vmin Vpeak Vripple VACin sin o t Vmin Since Vpeak is reached at the input sinusoid top or one fourth of the input period then the diode conducting time tc is simply in 1 Vmin sin m i z 3 360 Fline ic 1 Fine 3ms Vin 90 VAC During these 3 ms
9. line 1 6 A Maximum secondary RMS current 6 9 A Primary inductance 700 uH Turn ratio power section Np Ns 1 0 166 Turn ratio auxiliary section Np Naux 1 0 15 Clamping Network The clamping network can be calculated using the following formulae Rclamp 22 2 Vclamp Vclamp Vout Vf sec N Lleak Ip2 Fsw Vclamp 23 Cclamp Vripple Fsw Rclamp The power dissipated by Rclamp can also be expressed by 24 Vclamp Vout Vf sec N PRclamp 3 Leak Ip Few aes VoutrVfsec SN with Vclamp the desired clamping level Ip the maximum peak current e g during overload Vout Vf the regulated output voltage level the secondary diode voltage drop Lleak the primary leakage inductance N the Ns Np conversion ratio Fsw the switching frequency Vripple the clamping ripple could be around 20 V With a measured leakage inductance of 12 uH and a final clamping level of 150 V Rclamp is found to be 4 7 kQ 6 W and Cclamp 100 nF The RMS current flowing through Cclamp is 220 mA RC networks are economical clamping devices and care must be taken to not exceed the MOSFET BVdss in the most stringent conditions e g a cold startup sequence at high line Worse case arises when Ip is maximum and Vout reaches the target Stability Analysis The stability analysis can be investigated using different approaches Spice has proven to be rather accurate for feedback loop ana
10. or sustain life or for any other application in which the failure of the SCILLC product could create a situation where personal injury or death may occur Should Buyer purchase or use SCILLC products for any such unintended or unauthorized application Buyer shall indemnify and hold SCILLC and its officers employees subsidiaries affiliates and distributors harmless against all claims costs damages and expenses and reasonable attorney fees arising out of directly or indirectly any claim of personal injury or death associated with such unintended or unauthorized use even if such claim alleges that SCILLC was negligent regarding the design or manufacture of the part SCILLC is an Equal Opportunity Affirmative Action Employer This literature is subject to all applicable copyright laws and is not for resale in any manner PUBLICATION ORDERING INFORMATION LITERATURE FULFILLMENT N American Technical Support 800 282 9855 Toll Free ON Semiconductor Website www onsemi com Literature Distribution Center for ON Semiconductor USA Canada P O Box 5163 Denver Colorado 80217 USA Europe Middle East and Africa Technical Support Phone 303 675 2175 or 800 344 3860 Toll Free USA Canada Phone 421 33 790 2910 Fax 303 675 2176 or 800 344 3867 Toll Free USA Canada Japan Customer Focus Center Email orderlit onsemi com Phone 81 3 5817 1050 www BDTIC com ON m Order Literature http www onsemi com orderlit For additional information please cont
11. 00 we recommend a 18 KQ for R and 1 nF for C These values offer an acceptable tradeoff in terms of power consumption but also in terms of noise immunity The corresponding time constant of 18us gives a ramp maximum peak voltage of Vcc Vdrv 1 e7 5V 21 with t 0 45 x 1 61 k This provides an available ramp level of 677 mV us S By setting Radd2 to 1 kQ Radd1 can be computed using the following formula Raddi 1k 22 5 kQ __s S m for m 60 Final tweak gives an 18 k 2 resistor for Radd1 Component Constraints In this section we will see how Spice can help us to precisely determine all the component constraints and thus calculate the necessary amount of heatsink they need The simulation schematic we adopted is given on Figure 9 and shows the NCP1200 wired as recommended in the data sheet Please note that the simulation fixture has been simplified to allow faster simulation time For instance the TLA31 and its network usually hamper the simulation time to find out the right operating point A Zener diode and a resistor help finding it in a much quicker way www BD ff com ON NCP1203GEVB gt Vsec lout R3 XFMR X6 u mo Y n 200 m RATIO 0 166 MBR20100 400pH 10m 8 Vout wu Vout e Rload lripple2 Lleak D4 12 uH Ovinput MUR160 Vdrain R15 3 D lt Idrain 1 0k NCP1200 10 5 jo lt x5 SFH610A NCP1200 Fs 65k 14 1N4148 D3 Vsense 1N963 16 Rsense 0 33 VFB Fig
12. 50 VDC voltage rail Tricks exist to slightly reduce it like the half wave diode but you will only gain between 20 30 Power dissipation as stated above all the current consumed by the IC is seen through pin8 This is due to the self adaptive feature of the DSS Should the IC current move up or down the DSS duty cycle will automatically adjust to deliver it The controller current depends on the internal IC consumption but also on the type of MOSFET connected to the output It therefore important to assess the total current drawn from the HV rail and checks the right compatibility with the package type All details are given in the NCP1200 dedicated data sheet and the application note AND8023 D As a result the answer lies behind your design constraints If you would like to have a precise Over Current Protection OCP trip point while driving a moderate size MOSFET DSS can be a good choice provided low standby power is not an absolute necessity In our case we want to drive a large MOSFET for a better efficiency but we need to reach the lowest possible standby power We will thus adopt an auxiliary winding configuration to permanently disable the DSS Solutions to various combinations of these constraints are described in the application note Tips and Tricks for the NCP1200 document number AND8069 D Semiconductor Compon j 1 bl cation Order Number June 2012 Rev WWW e CO m EVBUM2124 D NCP1203GEVB Self Powering t
13. NCP1203GEVB A 70 W Low Standby Power Supply with the NCP120x Series Evaluation Board User s Manual Introduction The NCP1200 represents one of the cheapest solutions to build efficient and cost effective Switch Mode Power Supplies SMPS As this design example will show the part definition does not confine the component in low power applications only but it can actually be used in Flyback and Forward supplies for virtually any output power The below example depicts a universal mains 90 260 VAC power supply delivering 16 5 V 4 5 A Beside its ease of implementation the NCP1200 excels in true low standby power designs This application note details how an amazing standby power of less than 100 mW can be reached at high line with a nominal 70 W board DSS or Not DSS The Dynamic Self Supply DSS lets you directly drive MOSFETS from the high voltage rail This option brings you several advantages as stated below True overload detection with UC384X based systems the switching oscillations are stopped in case the Vcc line drops below a given Undervoltage Lockout level UVLO This principle considers a good coupling between the primary auxiliary winding and the power secondary winding Unfortunately leakage elements often degrade this coupling and you only can detect true short circuit when Vout is close to zero and not overload conditions Thanks to the DSS the NCP1200 does not need an auxiliary information to sense an
14. They however have all been tested okay Measurements were taken with the Coilcraft transformer As a final note the actual demoboard delivers 19 VDC 70 W versus the original design based 16 5 V As a result figure 1b circuit has been replaced by L3 R13 and C12 to improve the short circuit protection when using an auxiliary winding Output voltage can be adjusted by changing the feedback network made of R12 R20 R21 www BD ff com ON NCP1203GEVB lout X1x D1 NCP1200 XFMR MBR20100CT puti bt Rs T averaged 108 RATIO 0 166 163 10 uH m IN OUT 2 2 ee 16 9 16 8 CTRL FB GND 2 93 17 5 Resr1 R17 16 8 340 1 mcs oe c 30m 300 m Vin CoL NCP1200 Av 7 9 42 350 TURF FS 61k Oll 44 L 700u C1 C2 MC 39605 5 8 mF 100 uF VStim J RI 0 33 AC 1 out1 out2 x3 R15 SFH610A 1 0k Viii BEES 16 1 a R5 47k 15 4 C1 Rupp 0 400nF 939k C5 1 0 nF X5 2 50 13 TL431 Rlow 6 8 k Figure 11 The Simulation Schematic for Our 70 W Current Mode Power Supply Mag dB rr rn Phase deg 48 0 160 48 0 160 0 36 0 Mu 120 0 Phase 24 0 80 0 24 0 80 0 12 0 40 0 o Gain 0 0 12 0 40 0 24 0 80 0 24 0
15. Yes Resistor R3 Chip Resistor 1 0 MQ 0 25 W Vishay CRCW12061M0JN Yes R4 Chip Resistor 220 Q 0 25 W Vishay CRCW1206220RJN Yes R5 R8 Chip Resistor 10 kQ 0 25 W Vishay CRCW120610KJN Yes R20 R6 Chip Resistor 1 0 kQ 0 25 W Vishay CRCW12061K0JN Yes R11 R7A Chip Resistor 1 02 1 0W Vishay CRCW12181ROJN Yes B C R9 Leaded Resistor 47 Q 1 3 W Vishay CCF5047RFKE36 Yes R10 Chip Resistor 12 KQ 0 25 W Vishay CRCW120612KJN Yes R12 Leaded Resistor 27 KQ 1 3 W Vishay CCF5027KJKE36 Yes R13 Chip Resistor 1 5 kQ Vishay CRCW12061K5JN Yes R18 Leaded Resistor 1 0 KQ 1 3 W Vishay CCF501KOFKE36 Yes R19 Not Installed R21 Chip Resistor 5 6 kQ 0 25 W 1206 Vishay CRCW12065K6JN Yes C1 Polyester Film 100 nF 400 V pitch Vishay 2222 373 53104 Yes Capacitor 15 mm C2 Class X2 Suppression 470 nF X2 pitch Vishay F1772 447 20 Yes Capacitor 27 5 mm C3 Aluminum Snap In 220 uF 400 V Radial Vishay 2222 157 46221 Yes Capacitor C4 Aluminum Capacitor 100 uF 35 V Radial Panasonic ECA1VM101 Yes C5 7 Aluminum Capacitor 2200 uF 35 V Radial Vishay 2222 136 50222 Yes C8 Not Installed C9 Ceramic Chip 100 nF 16 V 1206 Vishay VJ1206Y104KXJAT Yes Capacitor C10 Ceramic Chip 1 0nF 16 V 1206 Vishay VJ1206Y102KXJAT Yes C23 Capacitor C11 Ceramic Chip 10 nF 16 V 1206 Vishay VJ1206Y103KXJAT Yes C12 Capacitor C22 Electrolytic Capacitor 470 uF 35 V Radial Panasonic ECA1VM471 Yes C24 Electrolytic Capacitor 47 uF 25 V Radial Panasonic ECEA1EN470U Yes C25 Cer
16. act your local Sales Representative
17. ail info coilcraft com http www coilcraft com ref Z9260 A with auxiliary winding P 70 W ref Z9007 B without auxiliary winding P 70 W ON Semiconductor and Q are registered trademarks of Semiconductor Components Industries LLC SCILLC SCILLC owns the rights to a number of patents trademarks copyrights trade secrets and other intellectual property A listing of SCILLC s product patent coverage may be accessed at www onsemi com site pdf Patent Marking pdf SCILLC reserves the right to make changes without further notice to any products herein SCILLC makes no warranty representation or guarantee regarding the suitability of its products for any particular purpose nor does SCILLC assume any liability arising out of the application or use of any product or circuit and specifically disclaims any and all liability including without limitation special consequential or incidental damages Typical parameters which may be provided in SCILLC data sheets and or specifications can and do vary in different applications and actual performance may vary over time All operating parameters including Typicals must be validated for each customer application by customer s technical experts SCILLC does not convey any license under its patent rights nor the rights of others SCILLC products are not designed intended or authorized for use as components in systems intended for surgical implant into the body or other applications intended to support
18. amic AC 2 2 nF 500 V Disc Vishay WKP222MCMBFOK Yes Capacitor Y1 B1 Diode Bridge 600 V 4 0A NA Vishay KBU4J Yes D1 20 A 100 V TO 2203 ON MBR20100CTG Yes Schottky Rectifier Semiconductor 1N4148 Diode 75 V 150 mA DO 35 Diodes Inc 1N4148DICT ND Yes D2 4 Digi Key D5A Zener Diode 27 V 500 mW SOD 122 ON MMSZ27T1G Yes Semiconductor Table 2 BILL OF MATERIALS NCP1203GEVB Not Installed www BD fI com ON Substi RoHS Desig Toler Foot Manufacturer Part tution Com nator QTY Description Value ance print Manufacturer Number Allowed pliant D6 1 Schottky Barrier 30 V 200 mA NA DO 35 ST 497 2495 1 ND Yes Yes Diode Microelectronics Digi Key D8 1 Ultrafast Rectifier 1A 600 V NA Axial ON MUR160G No Yes Semiconductor D9A 1 Zener Diode 15 V 500 mw NA SOD 123 ON MMSZ15T1G No Yes Semiconductor IC1 1 Optocoupler 5300 Vrms NA SMD 4 Vishay SFH6156 2 Yes Yes IC2 1 Programmable 2 296 TO 92 ON TL431ILPG No Yes Precision Reference 2 5 36 V Semiconductor IC3 1 Flyback BWM NA PDIP ON NCP1200P60G No Yes Controller NA Semiconductor Q1 1 PNP Silicon Plastic NA TO 92 ON P2N2907AG No Yes Transistor NA Semiconductor Q2 1 NPN Amplifier NA TO 92 ON P2N2222AG No Yes Transistor NA Semiconductor L1 1 Inductor 10uH 5A 1096 NA Coilcraft PCV 2 103 05 Yes Yes L2 1 Current Compensated 2x27 mH 30 NA EPCOS B82724 A2142 N1 Yes Yes Double Choke 1 4A L3 1 Molded Axial Inductor 47 uH 110 mA 1096 Axial Vi
19. eatsink according to the following calculation _ Timax Tamb max P R heatsink air R Junction Case R Case Heatsink 15 C W Lower R9 heatsink air resistances can of course be selected to run the device cooler Diode The MBR20100 welcomes two diodes that share nearly equal current thanks to their equal forward drops The total forward drop dissipation will remain the same but the RMS losses sensitive to the dynamic resistance will divide by two Irms total 6 8 A IAvG total Iout 4 2 A Rd 3 4 Ar 27 mQ Vf 2 2 Aayg 0 7 V Pcond for one diode 3 42 x 0 027 2 2 x 0 7 1 85 W or 3 7 W for the whole TO 220 package Simulations gives a bit less to 3 4 W Heat calculations Tj lt 100 C and 50 C ambient recommend a heatsink of 8 C W for the MBR20100 As stated before lower ROheatsink air resistances can of course be selected to run the device cooler Capacitors Icapacitor RMS 5 A The paralleling of capacitors will help achieve the right ripple current shared between all the devices We selected three 2 2 mF capacitors capable of handling 1 7 Arms each www BD ff com ON NCP1203GEVB Transformer Below are the key parameters you will pass to your transformer manufacturer to help him select the right winding size and tailor the internal gap Maximum peak primary current including 160 ns propagation delay 1 0 33 374 x 160 n 700 u 3 2 A Maximum primary RMS current at low
20. he Controller in Standby An auxiliary winding does not usually cause any self supply problem with a continuous pulses flow In standby whatever implemented frequency reduction techniques e g skip or frequency foldback the recurrence between pulses can become very low By definition the feedback loop manages to keep the energy content in each burst high enough to maintain the nominal output voltage However on the auxiliary side it can be difficult to keep the Vcc above the controllers UVLO Remember to permanently disable the DSS you need to guarantee a level above VccON max which is 11 V for the NCP1200 Failure to do this will re activate the DSS in no load conditions and standby power will be degraded Figure 2 offers a view of a typical bunch of pulses captured in standby at a 127 VDC input voltage ENDE IT i FONT TEE Figure 2 A Bunch of Auxiliary Pulses Captured While the Supply Operates at No Load Vin 127 VDC Rlimit 1N4148 Figure 3 The Auxiliary Is Clamped to Avoid Exceeding the 16 V Maximum Rating As we previously stated we want to deliver 70 W with a 16 5 V output level The maximum rating for the NCP1200 states a level less than 16 V As a result the auxiliary Vcc shall be less than 16V but also above VccON in any conditions to ensure full DSS de activation A solution consists in artificially raising the ratio between the power winding and the auxiliary one to ensure adequate suppl
21. he transformer parameters electrical but also physical ones including wire type bobbin stack up etc b you only evaluate the electrical data and leave the rest of the process to a transformer manufacturer We will adhere to the latest option by providing you with a list of potential transformer manufacturers you can use for prototyping and manufacturing However as you will discover designing a transformer for SMPS is an iterative process once you freeze some numbers it is likely that they finally appear either over or under estimated As a result you re start with new values and see if they finally fit your needs To help you speed up the transformer design a design aid spreadsheet is available from the ON Semiconductor web site www onsemi com pub NCP1200 Let s start the process with the turn ratio calculation Turn Ratio and Output Diode Selection The primary secondary turn ratio affects several parameters The drain plateau voltage during the OFF time the lowest plateau gives room for the leakage inductance spike before reaching the MOSFET s BVdss Np 10 Vplateau Ns Vout Vf VinDC max The secondary Peak Inverse Voltage PIV is linked to the turn ratio and the regulated output voltage by PIV pe VinDC max Vout x If you lower the plateau voltage you will increase the reverse voltage the secondary diode must sustain With these numbers in mind you can tweak the turn ratio according to the MOSFET
22. ly internal ramp compensation On NCP1216 and 1217 the internal ramp compensation avoids using the external circuitry made of RI R7 D4 and C10 If one of these two parts are plugged in the demoboard you must disable this network by simply disconnecting R1 and growing R6 up to 2 7 KQ for a 65 kHz operation The NCP120X takes place with two other bipolars that implement a discrete SCR activated in presence of an OVP e g an optocoupler failure D5 senses the overvoltage condition and can easily be adjusted to fit any other levels Thanks to R10 the OVP permanently latches off the supply and the user shall cycle Vcc off and on again to restart the supply Shutdown is obtained by pulling the feedback pin down through D6 The clamp resistor is split in two different components to avoid an excessive heat burden on one single device Both main MOSFET and secondary diode are mounted on an adequate heatsink to evacuate the heat To ease the designer task or simply help evaluating the board performance faster we have experimented different transformers available through Appendix B manufacturers Please note that some include the auxiliary winding for DSS deactivation whereas other only offer a dual winding arrangement where the DSS no longer activates and offers the best standby power All details are given in appendix B The final demoboard will not accommodate with all these transformers simply because multiple footprints was not possible
23. lysis with SMPS We will use the NCP1200 average model which is available to download from our web site www onsemi com pub ncp1200 Figure 11 shows the simulation template where the feedback network on the TL431 has been simplified to a simple 100nF capacitor Thanks to average modeling the simulation time is kept short and results are delivered in a snap shot as testified by Figure 12 Figure 13 unveils the results obtained using a network analyzer and confirms the validity of our approach Vin 240 VAC Stability has been checked at various line loads combinations and gave good results Final transient step did not reveal any overshoot or unwanted oscillations The Adapter Schematic The final schematic implements a current mode Flyback architecture driving a 600 V MOSFET The 2SK2545 features a 10 A capability but a 6 0 A 600 V can also be mounted such as the FOP6N60 from Fairchild but to the expense of increased conduction losses Figure 14 offers a complete view of the electrical sketch The board can actually be used with either auxiliary or without auxiliary winding By removing the resistance R4 you reactivate the DSS on a NCP120X controller featuring this ability The board can therefore accept the following controllers NCP1200 featuring DSS NCP1200A featuring DSS NCP1203 auxiliary winding only NCP1216 featuring DSS and internal ramp compensation Improved EMI jittering with DSS NCP1217 auxiliary winding on
24. nd goes up to 375 V The reflected voltage pushes further up by 101 V Summing up these levels gives a total steady state drain voltage of 476 V The last term in equation 18 depicts the leakage inductance action which further stresses the MOSFET at the opening If we select a 600 V device it leaves more than 100 V for this leakage action A clamping network will stop its rise anyway A 2SK2843 from Toshiba can be a good choice This is a TO 220 600 V 10 A component which features a 1 2 Q RDS ow Tj 100 C Ramp Compensation With a supply entering CCM together with a duty cycle greater than 50 we need to inject ramp compensation into the controller to prevent subharmonic oscillations An easy way to generate a ramp is to take the driving signal available from pin5 and integrate it through a RC network Figure 7 shows how to wire the components and Figure 8 shows the signal obtain with a 18 KQ 1 nF RC time constant To calculate the necessary amount of ramp m several methods exist We will stick to the standard one which consists in injecting between 50 and 75 of the off time downslope The calculation is as follow Primary off slope 19 N Wout L 153 mA us 19 Once reflected over Rsense it becomes 50 5 mV us S Duty cycle in CCM Vout Ep T 20 D qvin Vout 45 Vin 120 VDC From Figure 7 network the maximum voltage is given by R and Radd1 Radd2 With a 11 V driving voltage delivered by the NCP12
25. or this reason a transition from Discontinuous Conduction Mode DCM to CCM will be envisaged here Figure 6 depicts these different modes www BD ff com ON NCP1203GEVB Different methods exist to find the point transition takes CCM place also called the critical or borderline point The idea consists in finding the critical inductance Lc that will make the supply enter CCM at 180 VAC From Figure 6 we can en write I la ton Lp Ip 14 Not 0 at Pd VinDC Ip turn ON A 0 ie Lp Ip 15 9 N Vout Vf IL avg H L eM 0 before turn ON i BCM 0 Bis Dead time time Figure 6 Depending on the Primary Current at Turn On the Supply Crosses Various Operating Modes DRV 4 CS Figure 7 A Very Simple Way to Generate a Ramp from a Square Wave Signal 2 R D 3 Radd1 2 NNV e 1 R sense EN E G Radd2 150 From the Flyback formula we obtain Ip 2 Pout yn Lp Fsw Ip primary peak current N Np Ns 1 0 166 6 Pout output power n efficiency Lp primary inductance Fsw switching frequency Vf secondary diode forward drop VinDC Vac 2 neglecting ripple Combining equations 14 15 and 16 we obtain an Lp value to be in BCM at 180 VAC input voltage Vout 2 Vout Vf Vf2 eff N2 Vin2 17 Lp Pout N Vout N Vf Vin
26. shay IMO2EB470K Yes Yes 1 TO 220N 2SK2543 Yes Yes M1 Power MOSFET 500 V 8A NA IS Toshiba 1 Z9260 AL or Yes T1 Flyback Transformer 0 7 mH 1096 NA Coilcraft Z9007 BL No Heat 1 Seifert sink 1 Heatsink SW diode NA NA NA Electronic KL194 38 1 Yes Yes Heat 1 Heatsink SW Seifert sink 2 MOSFET NA NA NA Electronic KL195 38 1 Yes Yes NCP1203GEVB Appendix B Transformer Manufacturers Eldor Corporation Headquarter Via Plinio 10 22030 Orsenigo Como Italia Tel 39 031 636 111 Fax 39 031 636 280 eldor eldor it www eldor it ref 2074 5059A no aux winding P 70 W Pulse Engineering Site d Orgelet Zone industrielle 39270 ORGELET Thomson Multimedia Orega Route de Noiron B P 24 70101 GRAY Cedex France Tel 33 0 3 84 64 54 26 Fax 33 0 3 84 65 18 45 www thomsonmultimedia com Email Bouillotj thmulti com Ref G7086 01 no aux winding P 70 W For Lower Volumes Atelier Special de Bobinage 125 cours Jean Jaur s 38130 ECHIROLLES France Tel 33 0 4 76 23 02 24 Fax 33 0 4 76 22 64 89 Email asb wanadoo fr Ref NCP1200 35 W UM no aux winding RM10 P 35 W Tel 33 0 3 84 35 04 04 Fax 33 0 3 84 25 46 41 http www pulseeng com Email vpelletier pulseeng com ref PF0082 with auxiliary winding P 50 W ref PF0091 without auxiliary winding P 50 W Coilcraft 1102 Silver Lake Road Cary Illinois 60013 USA Tel 847 639 6400 Fax 847 639 1469 Em
27. ure 9 The Simplified Simulation Schematic Helps to Determine All the Component Key Parameters www BD tf com ON NCP1203GEVB idrain VCLAMP Iprim Iripple1 RMS 1 10 amps between 2 05 M and 2 06 M secs RMS 1 53 amps between 2 05 M and 2 06 M secs RMS 4 65 amps between and 2 05 M secs y mean 152 volts x first 2 00 M secs 2 01 M 2 03 M 2 05 M 2 07 M 2 09 M Figure 10 Complete Simulation Results of the 70 W Converter Operated at 120 VDC Input Voltage Important results appear in Figure 10 Please note that the maximum RMS current occur at the lowest line where the duty cycle is pushed to the limit As you can see the ramp compensation works fine and no subharmonic oscillations can be noted Once everything is extracted below are summarized the most important design constraints MOSFET Rdson 100 C 1 2 ohms RthetaJC 2 8 C W Poond 21 2 1 1 1 5 W The conduction losses are the strongest at low line The total simulated losses including switching events are evaluated to be around 2 6 W Further breadboard measurements confirmed this number If we want to keep the junction temperature around 100 C at an ambient of 50 C then we shall add a proper h
28. y at no load We successfully tested a 0 9 ratio where the auxiliary output gets clamped by a 15V Zener diode in nominal operation Figure 3 shows the option We measured a Vcc of 11 5 V 230 VAC and 12 2 V 90 VAC Rlimit on Figure 3 can easily be adjusted to move these values up or down depending on the final winding ratios Care must be taken to avoid over dissipation of the 15 V Zener diode in nominal conditions Power Supply Element by Element Design Let s first detail the specs of our power supply Vin 90 265 VAC Vout 16 8 V 4 2 A Pout 70 W Short circuit protection Over voltage protection Efficiency gt 80 Pin 70 0 8 87 5 The below sequence details step by step the calculation procedure for every component of the power supply www BD ff com ON NCP1203GEVB DC High Voltage Rail From these above numbers we can deduce the level of the high voltage rail neglecting the dual Vf drop VHV max 265 2 374 VDC Vuy min 90 2 127 VDC Vripple Figure 4 A Typical Ripple Voltage Over the Bulk Capacitor Bulk Capacitor Figure 4 portrays the typical waveform captured across a bulk capacitor delivering power to a given charge To simplify the calculation we will neglect the charging period and thus consider a total discharge time equal to 1 2 Fline From the design characteristics we can evaluate the equivalent current Iload drawn by the charge at the lowest input line condition
Download Pdf Manuals
Related Search