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1. 1UF R33 10K ME GND GND X1 X2 GND GND 15V sS 3 4 ji 7 4 ci ae usa Y 1UF pad LF347 hy 100K 3 cw to 1 PHASE ADJ 20x 3 R35 oy cis REF LEVEL ADJ TPZ v eas fox 047UF T 11 15v U6B TPS ci3 1UF RB RI LF347 i om PANE 1 0K 4 99R BLO REF 7 0KHZ W T z 7 510 5 FROM FIGURE 11 c16 3 16 Taba FEN SASUE AFC MODULATION INPUT CW R32 10K MODULATION RANGE Figures 10 and 11 Schematic of circuit card 26 containing AFC and Smith Chart circuits and U11D Fig 11 provide impedance transforma tion signal gain and bandpass filtering to condition the relatively weak and distorted signal available from the crystal oscillator up to the level needed by the other circuits 8 V sine wave This circuit sup plies the 70 kHz reference for the detection DBM described above and also is sent out to the micro processor interface to supply the reference for the phase tuning meter described below Limiter Figure 5 shows protection limiters 76 and 98 These limiters provide protection for the low noise preamplifiers The reflection from an overcoupled res onator can be destructive to a preamplifier The lim iters are positioned to protect whichever preamplifier is selected The limiter Fig 12 is made from type 1N4148 fast switching diode
2. LP347 R5 i Hij Gx ANY p cw L c4 pi 100K a AMOUR 1N4148 1UF alt 12V 1UF R9 R11 1 0K 1 0K R7 1 0K vie LF347 u1lD OUTPUT TO KEPCO R10 LF347 ATE 36 30M rI 2 1 0K R6 142 gt 8 10 0K F 14 1K WA Ae AM cs T 1UF POWER SUPPLY CONNECTOR 127 fel saa ce 330F GND CT T 33UP 12V zaa Figure 29 Schematic diagram of Bruker to Kepco magnet power supply interface PULSED AND CONTINUOUS WAVE 250 MHz EPR SPECTROMETER 89 the signal from a LGR can be amplified only after it has passed through the circulator With high incident RF power the power reflected from a LGR could saturate one of the amplifier stages if maximum avail able amplification were used so several options are available Baseline drift is a potential problem in both CW and pulsed measurements so we commonly use less than the maximum gain one might expect to be able to use to avoid reaching the limit of some stage during long time averaging The detection system of the pulse bridge includes pulse amplifiers locally designed and constructed with selectable gains that have been interfaced to the SpecJet high speed digitizer in the Bruker E540 con sole which has 0 5 V full scale input dynamic range The SpecJet provides dual channel digitization and high speed averaging capability 6 Presently we are doing the high speed timing control with our locally designed PTU 54 56 wit
3. I OG 7 v veo 5 1 10 23 10 24V ero tE8Y l T 33pe cal 1 18 _pscanon U29A Jeaan 5v LF347 3 ces ji 2 T 1UF vrerse 3 R920R93 L nee a st SCAN VOLTAGE A FIELD OE ADS D3 Da3 paz RFBA Tal 15V k FOR BOP oural2 1uF gt OLD MAGNET agnpal4 gt TP29 ETUNA F ores 10V 1206 22 BEBE af U29B 23 gera LF347 ouTB LUF 24 6 AGNDB x 7 DGND amp cmb 12 ps Figure 21 the source FM ramp dip displays or by one of the Smith Chart quadrature signals coming from the bridge resonator tuning modes Switch U6 Fig 17 makes this selection Stage UID amplifies the Smith chart signal coming from the bridge before it is ap plied to the x axis of the scope The y axis lower portion of Fig 18 is selected by switches U8 and U9 to come from either the lock in output spectral dis play the mode sweep signal from the bridge dip displays or the other Smith Chart mixer that is in quadrature to the one supplying the x axis drive in the Smith Chart mode This signal and the mode sweep signal are amplified by stage U7D The z axis retrace blanking is derived from the ramp generator circuit U3A 8051 microprocessor and serial communica tions interface Figure 23 shows 80C31 micropro cessor U35 and support chips The 80C31 is one in the generic family of 8051 8 bit microprocessors made by Intel Maxim Philips and others U38 decodes chip selects for peripheral devices
4. 3 7 T4AC04 LABI H A7 17 Ea 2 D7 D7 3 4 LAB6 1A6 16 R114 7 p D6 200K LABS AS osins MWA 50 LAB4 4 A4 papa TOS E ai i sv s LAB3 4A3 pitta oPLS ie LaB2 2 A2 pallto 6 JJP z LABI H A1 10 wk 507 pijp VATE LAB0 gt AO 9 Do H no 7 E 7418123 R C CE RITE wrn24d we 4 7K 18 aol ot sao tdce sd Zh12 RDN OE NINT A ap GND 12 cL ai gt 5V Figure 23 modulation coils have a field constant of 7 5 G A so a 60 G p field sweep requires 8 A pp or 4 Vp at the input to the power amplifier Modulation Coils The modulation coils are of local design and con struction The coils currently in use are wound on machined plexiglas forms with an average winding diameter of 17 8 cm Each coil is 75 turns of no 14 AWG varnished magnet wire The total inductance of both coils in series and approximately 12 ft of twisted lead feed wire is 2 8 mH The measured field constant at the center for Helmholtz spacing is 7 52 G A Because of our large resonators we have to mount the coils at greater than Helmholtz spacing which results in 10 15 reduction in the field constant The resonators also have only fractional penetration so we have calibrated the modulation amplitude at the sample by empirical methods Bruker to Kepco Interface Circuit for Field Control The interface unit shown in Figs 2 and 4 between the Bruker field controller and the magnet power supply Kepco model ATE 36 30M is
5. 6 p 5 4 74LS123 L t LAB3 T A3 Pan ack roe as R C CE Tr4 ERROR LIGHT L c122 LAB2 A2 TXDA 2 13 T vioc FT Spt tasi 2 al eee n9232 3 warcupoc 4p Cr E o Z4ACO2 CR14 GREEN Laso H ao rxpaf C1488 oy di 5V 3 ERROR tess cag c90 pI 87 8 cL T ivr 10r T 1ur pet n6 19 E3 3 ps 22l a5 comz ovr ro U46 c92 sv palsa raft sae iaaea vances sex TAAGO R113 c91 C92 coat cad pel pS R117 5V 9 ai 220 a Ba Sue T ue Paes cent ese rae LOCKINERR alas xemo 11 10 p24 52 R122 Ay OY D as 10 i5 7 13 g D132 INTRN NINT 1K c w gjase xei os T43A 9 ces TARA g po Bo TUNEMODE Sjas 0 spi n4 U49 094 z 3 Pig Reset l RESCAP as ya D3 74AC573 2 T WRN Z WEN opo HILOVOLT a3 v3t 8 y2 siho aoii eben cou KA 2 RDN _jRDN OPI e es rswerup Ja2_ Y2 a D1 petrn a 13 oxwow R8232 P2 RXDA 1UF 1UF MODOVTMP Al yi D0 7 14 c98 7 D5 6D 6g FSWPON 180PF AANE 1 pa 4sp sgh 8 opow U45A C90 gi ga 5 ol GND TAACO4 MOD OVERTERP A A D3 z 4o AIRCORE Faa 14 Horaris a pni dstr ve p2a 3p aq BRIDGEPWR Mc14a9 L y pi 72p 2g WAITINGCPU zi 4 6 4 css Do 1D WATCHDOG u4ac U45B S AN 7 MC1489 74AC04 R122RED LEN R8232 P2 c99 10 a 220 F 22 pi RXD WA U1lOB PEPE coms IN 9 zeae b cs6 i 88232 P2 e100 vase KRN 4 she FIELD SWEEP OVERTEMP T 1g0pF JAACOA AIRCORE 9 PsweTMp _ cRi2 GREEN LED Dt asa 10 BOON 17 wasn Cy LED 74AC04 R123 RED 9 p 8 1a MODON j 11 m
6. Iwr 74AC02 zapt U12B vi2zc SEUR SSS CMOnE ui2D T4ACO4 7aacoa TACOS 7 R33 3 4 5 TUNEMODE 2 s 4 3K Figure 18 U55A subtracts the peak detected DC voltage repre senting the measured modulation current from the command value to produce an error voltage This error voltage is applied to variable gain stages U24A and U24B Fig 20 upper right These stages amplify a sample of the sinusoidal modulation frequency be ing supplied from the HP 3310B function generator This sinusoid comes into stage U55B as 1 Vip the level required to provide the lock in amplifier refer ence It is then attenuated by a resistive ladder and selected by U23A inversely in proportion to the mod ulation command exponent Amplifiers U24A and U24B amplify the sinusoid up to the level necessary to force the error voltage to near zero Very little steady state error is required because the transfer function of U24A and U24B is linear in decibels or exponential voltage so the loop gain is very high The output of U24B then goes through a scaling and impedance matching stage U25C and then out to the modulation power amplifier see section below Thus the loop is closed between a modulation command coming from the user via the PC and the measured modulation current The difference or error is forced to near zero The loop response time constant is set by the value of the low pass filter formed by R76 and C49 This time constant is 3 6 s or an equivalent low pa
7. 0 20 dB continuous Weinschel 910 20 11 55 Power amplifier 400 W peak 56 dB gain TTL blanking CPC 8T400 56 High power directional coupler 50 dB Werlatone C6117 57 Isolator Channel Microwave AU369 225 275 MHz 62 93 Video amplifier selectable gain DC coupled DC offset DU 17023A 64 Quadraphase phase shifter Pulsar model MO B2 412 65 GaAs switch absorptive SPDT Mini Circuits p n ZASWA 2 50DR 66 71 High pass filter Mini Circuits SHP 100 67 Crossed diode noise blanking circuit DU 17012 68 Detector diode Alan Industries 50D 1 70 97 GaAs switch absorptive Mini Circuits ZYSWA 2 50DR 72 Adjustable attenuator Alan Industries 50CAL10 0 10 dB 74 Directional coupler cryogenic nonmagnetic 20 dB 75 Low noise preamplifier cryogenic Berkshire Technologies U 250 2 76 Passive limiter for detector protection begins to limit at 6 dBm DU 17013 79 80 Det diode zero bias Advanced Control Components ACTP 1629PC3 83 Isolator Channel Microwave AU 369 225 275 MHz 85 94 Scope switching circuit DU 17025 92 Hybrid 90 splitter Synergy Microwave DQK 705S 96 Fixed attenuator 6 dB 97 X10 amplifier DU 17035 98 Passive limiter with sensing circuit DU 17013A 101 10 dB attenuator 102 Amplifier 19 dB gain NF 3 8 dB Mini Circuits ZFL SOOHLNB 103 Directional coupler 10 dB Mini Circuits ZFDC 10 1 Component numbers correspond to reference numbers in Figs
8. configuration for mode 4 the Bruker pulse mode In this mode the E540 console is used to control the field and data are digitized using the Bruker Spec Jet In this mode the PTU provides all of the timing control including a trigger to the SpecJet Because this is the only timing interface between the PTU and the Bruker console this mode is limited to FID experiments and fixed pulse sequence ESE Even tually we plan to control the timing aspects of the experiment from the PatternJet unit in the Bruker console and this will permit collection of stepped pulse ESE data into the Bruker As in the DU pulse mode the bridge can be configured for either an LGR or CLR resonator 66 QUINE ET AL POWER AMPLIFIER SPECJET y PULSE AUXILIARY UNIT g z Q FLUKE RF gt BRIDGE TRIG GENERATOR EXTERNAL PREAMPLIFIER X Y DISPLAY MAIN FOR MAGNET TUNE corts LGR age PROGRAMMABLE Q SPOIL A prerrrzer BRUKER E540 CONSOLE TIMING UNIT HA PROBE SENSOR PENTIUM PC LL MAGNET POWER SUPPLY INTERFACE CIRCUIT Figure 4 Block diagram of Bruker pulse mode Detailed Description of Spectrometer Components Bridge General The overall block diagram of the pulse and CW bridge is shown in Figs 5
9. portions of circuit card 26 70 kHz oscillator 25 and deviation meter 36 Reference num bers refer to Fig 5 The AFC system uses a separate crystal detector rather than the DBM EPR detector so that AFC can be established without regard to RF phase This is important because the AFC signal and absorption EPR are in quadrature to one another and therefore the best phase for absorption EPR is the worst phase for AFC The preamplifier provides low noise amplification of the detected 70 kHz signal with 40 dB 100X gain This is followed by the portion of circuit card 26 shown in Fig 10 In this diagram stages Ul U2 and U3 provide a gain selectable amplifier the gain of which is selected on the front panel of the bridge Three gain selections are avail able which are preset by pots on the circuit card Stage U4 is a moderate Q 20 70 kHz bandpass tuned stage followed by additional gain in stage U6A This stage is followed by double balanced mixer DBM1 which synchronously detects the signal The reference side of the DBM is supplied through stage U6B that has pots to set the phase and amplitude of the reference The output of DBM1 is then integrated by stage USA The output of the integrator is a DC error voltage that is low pass filtered by R33 and C25 low pass corner at 16 Hz This error voltage is then combined with the 70 kHz alternating current AC component in stage USC which is then filtered and buffered and sent to the DC FM po
10. through the microprocessor interface unit on its way to the lock in amplifier or Bruker signal channel This sample contains both the EPR signal at the field modulation lock in reference frequency and a com ponent at the 70 kHz AFC modulation frequency Since the EPR component is derived from an ampli tude modulation AM component at the detecting mixer in the bridge and the AFC is an FM compo nent they are in quadrature to one another This allows setting the proper RF phase for absorption EPR without being able to observe an EPR signal If the AFC component is nulled this will be the best phase for absorption EPR 58 Stage U7B has a gain of 6 This signal is then further scaled by a microprocessor controlled multiplying D A converter stages U13B and U15 The output of this stage is synchronously detected by mixer U34 Fig 22 The 70 kHz refer ence comes from the bridge via connector J9 and is scaled by stage U33A Stage U33D following the mixer provides a low pass filter corner frequency of 1 6 Hz and a low impedance source for the next stage Stages U33B and U33C constitute an absolute value amplifier which then drives a front panel null meter When the meter shows zero deflection the phase is properly tuned Zener diode CR8 prevents overdrive of the meter when the phase is improperly tuned This is a very sensitive and useful addition to the means of properly adjusting a CW EPR spectrom eter x y display drivers an
11. 183 Alecci M Gualtieri G Sotgiu A Testa T Varoli V Multipolar magnet for low frequency ESR imaging Meas Sci Technol 1991 2 32 37 Alecci M Penna SD Sotgiu A Test A R F 280 MHz EPR imaging of extended samples apparatus and preliminary results Appl Magn Reson 1992 3 909 915 Alecci M Penna SD Sotgiu A Testa L Vannucci I Electron paramagnetic resonance spectrometer for three dimensional in vivo imaging at very low fre quency Rev Sci Instrum 1992 63 4263 4270 Sotgiu A Alecci M Brivati J Placidi G Testa L New experimental modalities of low frequency electron paramagnetic resonance imaging In Ohya Nishiguchi H Packer L editors Bioradicals detected by ESR spec troscopy Basel Birkhauser Verlag 1995 p 69 92 46 47 48 49 50 Sh 52 53 PULSED AND CONTINUOUS WAVE 250 MHz EPR SPECTROMETER 91 Quine RW Eaton GR Eaton SS Pulsed EPR spectrom eter Rev Sci Instrum 1987 58 1709 1724 Quine RW Rinard GA Ghim BT Eaton SS Eaton GR A 1 2 GHz pulsed and continuous wave electron para magnetic resonance spectrometer Rev Sci Instrum 1996 67 2514 2527 Rinard GA Quine RW Harbridge JR Song R Eaton GR Eaton SS Frequency dependence of EPR signal to noise J Magn Reson 1999 140 218 227 Rinard GA Quine RW Eaton SS Eaton GR Fre quency dependence of EPR signal intensity 248 MHz to 1 4 GHz J Magn Reson 2002 154 80 84 Froncisz W Camenisch TG Ratke JJ Hyde JS Pul
12. 5 through 8 and Fig 13 NF noise figure PULSED AND CONTINUOUS WAVE 250 MHz EPR SPECTROMETER 63 FLUKE gt BRIDGE PHASE TUNE METER SR844 OR SR245 RF GENERATOR APC MODULATION COILS MAIN MAGNET COILS LGR SR510 LOCK IN AMPLIFIER x DISPLAY DIGITIZER PENTIUM PE HP 3310B8 MODULATION SOURCE MICROPROCESSOR MODULATION AND SWEEP AMPLIFIER MAGNET POWER SUPPLY INTERFACE UNIT Figure 1 Block diagram of DU CW mode in a Bruker SpecJet or sample an echo signal with a boxcar averager In the spectrometer described in this paper the EPR signal is generated in a resonator and after transformation and coupling to the transmission line as described in detail in previous papers from our laboratory 48 51 52 the S N is degraded by all losses and by the noise figures of all active devices e g amplifiers in the signal path In addition source noise reflected from a reflection resonator combines with the signal to reduce the S N below that inherent in the thermal noise of the resonator and sample Consequently the selection of a CLR is a fundamen tal design philosophy The CLR isolates the EPR signal from source noise and decreases the dead time in pulsed EPR as
13. R3 a i 34er J RI Py le VA sox vV GND UP RATE isv FST FIGURE 18 6 8200pE omp R6 47 5K A fe CR1 B RX 10K S VAN VWW SCA L cs 1 UP z T 6200pE 5V l 15V 5V RX al 4 7K V vec BRIDGE SAWTOOTH DG419 R17 20Vpp LK 4 99X Nt WW BST a l LS runemone At 74ac74 z i V GND 7 hB 15V sare SCOPE Z 12E R19 HORIZ SIZE LER R20 8 18 SMITH CHART A FROM BRIDGE Figures 17 through 26 Schematic diagram of microprocessor controlled interface unit circuit card containing circuits for linear sweep generation 70 kHz null meter field modulation control slow field scan control center field control microprocessor and support chips and serial commu nications interface modulation coils must be resonated as described in the sections on CW mode because the inductance of the large coils is so high but the DU modulation current control system uses the closed loop calibration An open loop system would be used if the swept fre quency specification were removed The modulation system consists of the control sys tem described here plus the power amplifier and mod ulation coils described below The modulation control is a part
14. T 7 15 33p war auo L5 i 4 R8 z5 100UK ji a gt T ae 100 9 i ANA t war vie L6 H gt E4 1000H 6 i y c25 50 wn 4 Bin t MET GRN gt capt c 1 c20 c19 c18 T 1UF T 1uF T 10r T 1uF Vv GND R7 4 5 40 2 T 1sv C10 OMIT E 2p sv E i Hy ees exon OL eer ye tPF gii ree 5V 3 5V R6 a En aL c12 a D 10 0 a T 10r LA z a a E a jd L v2 Yt is m CLC425 1 0K guce25 s onaran 2 RX3 INPUT 2 csa cop 6 1K 7 6 aa 6 Fi 3 3 r J1 3 qt f AM eee 2 dg 33UF 33UF R v v R12 Rp v R4 i P a DIS 49 9 8 4 RX2 49 9K z F sv 49 9K Ria ee aaa cit 332 Lcr L cs TF 1uF ka 10F 5V 1UF J 3 cxi 100UH TF 4 7uF wWH YEL f 1E VIN p c OFFSET ig 15V 5V L1 15V zi 1000H g L LM7905UP N OUT YEL Tee Let Lez SRP Lei Tes T 1uF 33UF T 1UF T 1UF I AN d Figure 14 Schematic diagram of video amplifier then using the DC offset control to set the DC position of the baseline The DC offset is injected into the circuit after the AC coupling The amplifiers Fig 14 are built around the CLC425 op amp which to our knowledge has the lowest noise of any available con sistent with the gain bandwidth product that is re quired for this application There are two gain stages followed by a cable driving stage The bandwidth of these amplifiers ranges from 45 MHz at a gain of 50 to 8 MHz at a gain of 500 The digital control signals that
15. a small amount of harmonic distortion but the resonator is an effective filter that eliminates reflects the harmonics The diodes have finite capacitance in the off state so the residual noise of the amplifier is not completely blanked However our tests indicate that this circuit pushes the noise down by 10 dB and we have ordered a power amplifier with internal blanking specified at 10 dB above thermal so this circuit should bring the noise essentially down to thermal level The diagram also shows quarter wave stub tuned short open branches These are designed to provide a short branch at the main line for the low power noise condition and an open branch during the power pulse thus further PULSED AND CONTINUOUS WAVE 250 MHz EPR SPECTROMETER 77 D17 D31 D45 mn t 1N4148 1N4148 1N4148 D18 D32 D46 J6 ou aaa be gt t gt t _ _ a 1N4148 1N4148 1N4148 pN D19 D33 D47 J T t 3 p M H 1N4148 1n4148 1N4148 i D20 D34 D48 D SR 4 m gt OA LINE 1N4148 1N4148 1N4148 LINE 4 D21 D35 D49 i 4 i tt 4 pit 1N4148 1N4148 1N4148 i D22 D36 D50 i 33 Pos men Mm as 1N4148 1N4148 1N4148 D23 D37 D51 Ht 4 4 D1 D2 D3 D4 1N4148 1N4148 in4148 Dg D10 D11 D12 F 1N4148 1N4148 1N4148 1N4148 D24 D38 D52 anana F 1N4148 zg y gt p zi 1N4148 1N4148 1N4148 DS D6 D7 D8 D25 D39 D53 Dis D16 X 1N4148 1N4148 1N4148 1n4148 t
16. bring in the gain control signals are extensively decoupled to reduce noise Also the power supplies are reregulated on the circuit card to reduce power supply induced noise Integrated circuit based ampli fiers with high gain bandwidth product are notori ously noisy We have measured noise figures for amplifiers 62 and 93 of 10 dB but because of the large amount of gain in the system before these am plifiers they do not contribute materially to the overall noise floor of the spectrometer The amplifiers are constructed on a small printed circuit card 6 86 cm X 4 32 cm of local design and housed in small boxes with SMA connectors Signal Distribution and Filters Following the video amplifiers in Fig 8 are the scope switching cards 85 and 94 These cards Fig 15 provide front panel selectable filters function selection and signal distri bution to the digitizer boxcar averager and monitor ing oscilloscope The front panel filter selections are 1 MHz 5 MHz or NONE Signal functions that can be selected at the front panel are incident pulse monitor derived near the output of the power amplifier from crystal detector 68 Fig 13 reflected pulse display see description under the Resonator Tuning Sys tem above and the time domain EPR signal These cards make it possible to observe the spectrometer output signals on an oscilloscope at the same time that they are also routed to either the SpecJet digitizer or to the boxcar
17. described in Ref 40 The isolation provided by the CLR also permits placing the low noise signal amplifier close to the output of the reso nator which cannot be done with a reflection resona tor Many of the modules of this system were designed and constructed locally that is in the laboratories of the University of Denver A complete documentation package including all engineering figures and sche matics plus additional detailed drawings not included in this paper is available from the authors Major Operational Modes Various combinations of the subsystems are possible which could result in many different operational modes The four major operational modes that will be described here are 1 DU CW 2 Bruker CW 3 DU pulse 4 Bruker pulse DU CW mode Figure 1 shows the system compo nents used in mode 1 the DU CW mode In this mode the resonator is most commonly a reflection resonator LGR but a CLR 36 40 can also be used The magnet used can be either the 81 cm or the 40 cm air core magnet If the 81 cm magnet is used it is powered by the Kepco power supply and if the 40 cm magnet is used it is powered by an HP model 6010A power supply Either supply can be controlled by personal computer PC software via the microproces sor interface box A complete description of the larger magnet and a brief description of the smaller magnet appear elsewhere in this issue Each of the major subsystems is described below In this
18. dip display can be observed from DBM 14 Fig 8 In TUNE the output of this mixer is amplified by ampli fier 21 and routed to circuit card 26 where it is amplified further and provided with DC offset From here it is routed to the microprocessor control box where it is ultimately sent to the y axis of the x y oscilloscope display We have found this dip display from the DBM useful in adjusting the length of the reference arm delay line since it preserves phase in formation in addition to the amplitude response Reflected Pulse Display When a resonator is adjusted for moderate to extreme overcoupling the Smith Chart becomes less helpful for either selecting the proper operating frequency or for estimating the resonator Q The bridge has a facility for observing the reflected response to an RF pulse which facilitates a judgment of the proper fre quency setting even for extreme overcoupling see Fig 7 in Ref 5 The reflected response also pro vides a means of measuring the resonator Q by mea suring the ring down time at the end of the pulse The reflected signal is obtained through circulator 6 LGR mode or through couplers 49 and 74 CLR mode and detected by crystal 79 Fig 5 By using a crystal and not a DBM the power reflected can be measured independent of the RF phase information The crystal is in the power law detection region below 10 15 mV We often digitize these signals so that we can obtain computer assisted curve fi
19. found that we could run the mod ulation system unresonated and obtain useful modu lation amplitudes in the 1 10 kHz range As noted in the prior section these low modulation frequencies are needed for very narrow line spectra such as the trityl radicals 53 For a specific modulation field goal coils could be designed to match the Bruker system DU pulse mode Figure 3 shows the system compo nents configured for mode 3 the DU pulse mode In this mode the field can be controlled from the PC using locally written software Optionally the field can be controlled by the Bruker console if the 81 cm magnet is in use The bridge can be configured for either a reflection resonator or the CLR Best results are generally obtained using the CLR due to the isolation between the source and the detected signal In addition the Q spoiling circuits used with the CLR substantially reduce the measurement dead time as described in another article in this issue 40 In this mode the locally designed programmable timing unit PTU 54 56 controlled by the PC provides all of the timing functions necessary to do the various stan dard two and three pulse ESE experiments as well as more advanced experiments involving complex pulse sequences Data are collected into a locally designed boxcar averager and digitized by a locally constructed digitizing box utilizing the Stanford Research model SR245 digitizer Bruker pulse mode Figure 4 shows the system
20. of the microprocessor based interface unit Modulation control begins in Fig 19 Stage U17 is a Hall effect sensing device through which the modu lation coil current is routed The output of this device is a voltage proportional to the current This signal contains both the sinusoidal modulation component and the linear ramp component 10 Hz Stages U18A and UI8C are a 3 kHz high pass filter that attenuates the linear ramp component This was de signed at the time in which the minimum modulation frequency was to be 5 kHz Stage U19 is the calibra tion and overall gain stage Pot R44 is mounted on the front panel so that a calibration adjustment can be made to account for the modulation penetration into our large metal resonators Note here that higher gain in this stage results in lower modulation amplitude This is because this path is the negative feedback path of the control loop Stages U20 and U22 are gain stages whose gain is set via the PC The modulation amplitude is controlled in three ranges 0 01 0 1 0 1 1 and 1 10 G The gains of these stages are set depending on which range the modulation amplitude requested by the user falls into These three gain ranges effectively constitute the exponent of the de sired modulation amplitude The output of stage U22 goes to Fig 26 which shows a full wave balanced rectifier and peak detector This circuit produces a DC voltage proportional to the peak to peak modulation current At t
21. panel controls are mounted The card is arranged so that some connectors protrude through the front panel and some through the rear panel of the box Modulation Control System The modulation control system contains design features that permit many modes of operation One design specification was to be able to control a calibrated current in relatively small low inductance modulation coils over a broad band of frequencies 5 200 kHz which could be swept to discover resonator microphonic resonances in small resonators This required that the modulation coils not be resonated and that the modulation power amplifier be capable of driving the coils directly This also required that the current control system operate with a current feedback element to maintain magnetic field modulation calibration at all operating frequen cies This is in contrast to fixed frequency designs such as the Varian E line system which is a resonated open loop calibrated design Later systems with more integrated computer control such as the Bruker system are able to do sophisticated computer controlled cal ibration at a number of frequencies but they still operate open loop For the specific large VHF reso nators and large modulation coils used to date the 78 QUINE ET AL 10Vpp R4 s osr R2 vz ls LAW Vt x W vec 5VAN DOWN RATE TP1 g R5 47 5K ANY 10HZ one ae 300 Hz LOW PASS v1B LF347
22. required to match the transfer function of the power supply and magnet to the Bruker The Bruker field controller was designed to control a power supply with a large gain and driving an iron core magnet with a large time con stant The interface circuit Fig 29 provides the gain and time constant required to stabilize the control loop Stage UIA provides gain of 10 and R4 R5 and C4 provide the time constant of 30 s UIC provides a summing junction to bring in a rapid scan control voltage that bypasses the large time constant This is useful to scan the magnet rapidly in certain types of experiments System Power Supplies In a system this large and complex a rather large number of different power supply voltages are re PULSED AND CONTINUOUS WAVE 250 MHz EPR SPECTROMETER 85 cea io 5 MC1488 100K E 2 3 5V ae SeN26a1 TAR gt 20 com 1 i cos COM3 GUT 7 1UF wi x 23 U44B R8232 P3 H T t O FP 2 6864 muz za x2 4 MCLsss a a 47A C93 ao F xz VART
23. such as D A converters and the communications port U40 is a ultraviolet erasable EPROM that contains the microprocessor object code RAM chip U41 provides for temporary storage of variables but this socket has not been used yet because the on board RAM of the microprocessor proved sufficient The source code was written in 8051 Assembler language and initially tested and developed using Signum Systems model USP 51 in circuit emulator The source code is 550 lines of 8051 assembly language code available from the authors The microprocessor receives instructions and parameters over a serial communications port from the PC at 9600 baud The protocol for this communication is defined in a document available from the authors UART chip U42 Fig 24 receives control parameters from and sends status parameters to the PC The UART generates interrupts to the microprocessor to process incoming and outgoing data bytes One shot U47B Fig 23 blinks a front panel light indicating communications activity The microprocessor runs at 10 MHz formed by oscillator U50 and divider U3B Fig 24 PULSED AND CONTINUOUS WAVE 250 MHz EPR SPECTROMETER roe B AM 7 49K sll cw 10K FIELD SPAN ADJ co 150 R95 3 48K LM399 VOLT REF L c74 1uF L FREFV 10 0V N aif R100 1 28 TP32 AN c76 Eaa 70KHz REF Toena FROM BRIDGE u33a Y k a
24. through 8 Part num bers and manufacturers for the circled reference num bers are detailed in Table 1 The operational modes of the bridge are detailed in Table 2 A detailed bridge wiring schematic diagram is available The bridge is constructed with all connectors power supplies and control interfaces such that it can be plugged directly PIELD CONTROL EXTERNAL FREQUENCY 3 arc DELAY LINE SOURCE FM APPROX 22FT RGSS Q9 p per HIGH PASS nso 20dBm FILTER os N BNC BNC 20aa e gaa 322 a DISCRETE 218 Taen FRASE ae RD suxPTER Eg 10am MECHANICAL zc17035 15ap Q Pms SHIFTER xe RP 5dBm iee apa ceteris taggin adgas veren Grai 2 6dBm HERE 104Bal Qa Ge Q SEE PIGURE 8 aan PTET a a Gana asp 1O us 200B SW rrure BLOCK PREQuency COUNTER ners DG Be 2 16878 TUNING SWEEP METER 2 ETAL l SOURCE A AFC AND SMITH A aa orr o METER CHART CIRCUIT o ome CARD BC 17024 7h X 20 REZ Hrn 350 is Sea ssa Q 513
25. z slur L DMH sia x 13 F T s ve S2A a 2 TP20 6la3q PA Saet apeoo EA U24B R62 7 S4A E A1LO 14 c45 4 RRA DG409 cae Es Pres aort a2ut apeoo 2 _ MUX l aicu H 1UF R69 2L0 5v Jen ciLo ae 11 e14 ai BOON SS oe A20PL nee Go H ao0 OFF cat 8 azca 22 1 VSS GN F e2u0 a 1 c42 v 5 3 15 T posaur a GAT2 A c43 R64 5V gt TP22 syi 2 2 i T 1UF 5V np m we c4a viacda R70 v12F 4 33UF AG R56X 200K TARCOA CRS L Hono 2 10 one 100K REFV4 13 12 TO a72 gt Formerly C56 10K T 2N4248 RIA 5V 200K C47 c53 TP16 1UF p i l i ratsias 7 5 l4 TP22 R C CE RIT 1K ala NER cas rol Ta eN TO POWER AMP ai gee FNP 54 a TT gt TP17 r 5 cu R76 ah cas F 36 4K iur T iuF i a c50 15V co 5VAN FaU 4 R73 ussa V 2 0K LF347 FROM FIGURE 26 DC 7 R74 3 z 2 0K FROM FIGURE 21 oyp 1UF uo y Ti Les Chae re 7 4 70F pice Sarel gii Figure 20 is a 12 bit inverting and multiplying D A converter The output of U29A is a voltage scan scaled to the selected scan range with a scale factor of 0 1023 V G The maximum scan range is 200 G 100 G from center field or 10 23 V at the output of U29A Switch U30 turns the slow scan system on or off selectable by the user at the PC The scaled scan voltage is then summed with the center field control voltage in stage U27C 70 kHz phase tuning meter Stage U7B Fig 18 acquires a sample of the bridge output that is passed
26. 1 10K Figure 26 PULSED AND CONTINUOUS WAVE 250 MHz EPR SPECTROMETER 87 WH BLU J2 HILOVOLT 9 PIN D FEMALE agi 19 PIN MS STYLE REAR PANEL m APEX AMPLIFIER H S 1 OVERLOAD LED ret 42000F D asv gt VIOLET ve i 7 VIOLET neat Dens ah ey 75 Ea WHITE vB r W me VB MODULATION DRIVE J5 Coax BNC REAR Ly JL v our BLACK 6 1N5234 oo 28 CHASSIS GND NOTE ALL WIRES 20 AWG MINIMUM EXCEPT 16 AWG MINIMUM S FT L 5 PIN MS FEMALE gt AT END OF CABLE REAR PANEL 3 33 3 PIN MS STYLE REAR PANEL REVERSING SWITCH FRONT FSDRIVE 31 MH FROM FIGURE 28 REAR PANEL HOLE WITH GROMMET f t Figure 27 Schematic diagram of modulation power amplifier to the CLR as described above By extrapolation it can be estimated that 4 3 X 10 spins with the same relaxation time could be detected with S N 1 in 1 s We observed a slope in the baseline of field swept spectra and found that it was due to magnetic field interaction with the circulator and isolator in the bridge The circulator was shielded in a6 X 3 5 X 2 5 inch box constructed of MuMetal by Mu Shield Com pany Inc Goffstown NH drawing 2751 The isolator was shielded with a homemade steel shield The shielding of these comp
27. 150 11 Low pass filter 450 MHz Mini Circuits SLP 450 12 10 dB directional coupler Merrimac CRM 10 500 14 DBM Merrimac DMM 2 250 15 Mechanical phase shifter Sage 6709 17 90 180 phase shifter Vectronics DP620 225 67HS 18 DC block Inmet 8039 19 42 43 63 73 77 84 86 87 88 99 100 Coaxial switch 12 V Dow Key 401 2208 21 100 kHz amplifier DU 16817A 22 31 Detector diode zero bias Schottky Omniyig ODZ0501AR 23 Not used 24 70 kHz AFC preamp DU 16817C 25 70 kHz oscillator DU built from International Crystal TO 11 70 kHz 26 AFC and Smith Chart circuit card DU 17024 27 90 hybrid splitter Merrimac QHM 6 165 28 29 90 91 DBM Mini Circuits ZFM 3 30 82 89 3 dB 0 splitter Mini Circuits ZFSC 2 1 32 AFC amplifier 12 5 dB gain NF 2 3 dB Cougar AC540C 33 34 Transfer switch Dow Key 411C 2208 36 AFC deviation meter R amp R Instruments 288 112 EMEM 37 Not used 39 Panel meter 200 mV 3 digit LCD Acculex DP 650 40 High pass filter Mini Circuits SHP 150 41 Frequency counter prescaler DU 16878 47 48 95 50 Q load SMA 49 High power directional coupler 30 dB Werlatone C6137 51 78 81 Directional coupler 15 dB Mini Circuits ZEDC 15 2B 52 Amplifier 38 5 dB gain NF 2 3 dB max output 30 dBm Miteq AUP 1479 53 Step attenuator 0 60 dB 10 dB steps Weinschel 116A 60 66 54 Variable attenuator
28. 2 LGR PULSE DETECTION 7a A RORMAL OSCILLOSCOPE INCIDENT Ln REFLECTED FILTER 20 MHz L NONE 5 MHz VIDEO GAIN 100 200 50 500 Q 1 D C OFFSET 70 QUINE ET AL sary to approximately match the phase delay of the reference arm to that of the path to the resonator and back By matching these delays the detection mixers provide better phase noise rejection by keeping the noise in the reference path coherent with that in the signal path This is equivalent to saying that all fre quencies have equal delays or what is known as flat group delay response in communication systems The reference arm phase is front panel adjustable by me chanical phase adjuster 15 and discrete phase shifter 17 At this frequency the mechanical phase adjuster can only shift the phase a little more than 90 end to end The discrete phase shifter has four 90 steps so that a total of a bit over 360 of phase shift adjustment is possible The discrete phase shifter is also used in the Smith Chart tuning mode to chop the display see description under Resonator Tuning System be low Directional coupler 103 10 dB is used in addition to coupler 69 20 dB because an appropriate 30 dB coupler was not available at 250 MHz Frequency Counter The front panel of the bridge displays the source frequency to six digits of resolu tion and accuracy The time base standard for this display is a temperature compensated crystal oscill
29. A Pulsed and Continuous Wave 250 MHz Electron Paramagnetic Resonance Spectrometer RICHARD W QUINE GEORGE A RINARD SANDRA S EATON GARETH R EATON Department of Engineering and Department of Chemistry and Biochemistry University of Denver Denver Colorado 80208 2436 ABSTRACT A 250 MHz electron paramagnetic resonance EPR spectrometer was constructed to be an engineering test facility for in vivo EPR imaging of physiological samples and for protein structure determination Innovations relative to prior low frequency EPR spectrometers include a four coil air core magnet and gradient coils a crossed loop resonator dynamic Q switching to decrease dead time in pulsed EPR and a narrow band bridge based on circulators The automatic frequency control system uses a signal separate from the EPR signal to make the frequency control independent of the radiofrequency RF phase The design incorporates multiple excitation and signal paths to facilitate testing of a variety of resonators two magnets and both a locally built console described here and a Bruker console Plug in cards in the bridge facilitate using reflection or crossed loop resonators in continuous wave or pulsed EPR modes In the locally built console there is a microprocessor controlled interface unit to handle magnetic field modulation and scan tuning display and other functions 2002 Wiley Periodicals Inc Concepts in Magnetic Resonance Magn Reson Eng
30. Cae 12aB arn TA 30aBm OUT 1 sees a z Logan aoe AFC DEVIATION METER TSOLATOR eTe Me ose Passive BIT 4 x LIMITER REPLACED 2 01 za K oan arru 1dB STEPS a 17 Ps E2 ap aren 412 5an Go mr sre L 194B GAIN EC 17022 DRESSLER 758 250 A soas F 17 Bm SATURATED i zuz NuaLian Pors ae S ee oc L SWITH CHART OSCILLOSCOPE DISPLAY o aap DOUBLE BALANCED MIXER 2 x spuirtex DOUBLE BALANCED MIXER 2 90 pacae SPLITTER Ioan Figures 5 through 8 Block diagram of DU constructed VHF bridge ATTEN 9 PULSED AND CONTINUOUS WAVE 250 MHz EPR SPECTROMETER 67 CW LGR MODULE 17014 LGR SOME COMPONENTS SHOWN MORE THAN ONCE FOR CLARITY PULSE LGR MODULE i 17017 co B Figure 6 into a Varian E Line console in place of the Varian bridge The bridge is not equipped with motorized remote control features that are compatible with a Bruker bridge controller The description given here is the operation of the bridge when connected to locally designed and constructed console functions and controls Front Panel Controls Figure 9 is a representation of the bridge f
31. K REF70KHZ TO FIGURE 10 22K z7 Glee ASTAN eee p cua loo LE347 waon vlin TP12 eee LF347 LF347 147 70KHZ OUT E z 4 99K 99K o 8 13 i3 13 ii c42 12 F 12 Pa T OL5UF 7OKHZ FROM OSC k RSS S C CHOPPER FREQ ADJ 31E R58 ANY 5V Tess sox CW neue 022UF GIIA Te TE A 74HC74 5 4 R C CE 2 74L8123 45 1 viza 2 da aP veg gmo TTE 7T4HCO0 USA usc u14B 3 9 A p4EC7266 TAACOA 24128 10 3 5 6 4 R63 wise 1K o 24EC7266 5V 10 ery TTLSON DOUBLE SINGLE SW x1 TRON TUORE z GND DOUBLE u8sD R75 TAACOA gosn 21 e ar 158 4p Do R62 T 25 t 5V i sv 25 1K o oe we c44 c34 c35 c36 59 WA R60 i R76 330F 7 1UF 7 1UF T 10F 51 i 4 7K T re i 4 a F T r 7 i 1 1 0 2 23 TO S C ON OFF SWITCH 39 20 isv 33 40 l 15V RES VARACTOR BIAS OUTPUT m IMEG cas be 33 36 Sopr OENE ENASTE D7 VARACTOR ADJ INPUT L C37 l C47 30 1N4148 noua 3307F FROM CONSOLE SCOPE GND 1N4002 c46 L OF REFERENCE ARM pa 7 COAX SW D10 330F 32 af r 1naiss 15v 15v 1N4002 Figure 11 amplifier minimizes the signal losses p
32. LED U45E 72 74ac04 MOD V HI LO gt HILovort t 10 cR13 GREEN Fswpon i3 X L LED u36 kH P T C a WAITCPU vase Ea z F 74AC04 RI24RED WAITINGCPU R10 GREEN 13 Moz 248 FscaNon i5 i LED U36D ak 74AC04 R125REH Dot 220 Figure 24 quired A total of 12 different DC voltages were required ranging from 48 V for the modulation power amplifier to 5 V for some op amps Another requirement was to make the bridge compatible with Varian E Line X band bridges which operate on f120 V The power supplies were constructed from commercially available units and packaged into larger boxes with alternating current power fuses and on off switches RESULTS AND DISCUSSION In the following paragraphs we comment on experi ence to date with the RF source the resonator the magnet and CW and pulse operation Other than the bridge and resonator there are two complete spec trometers one locally constructed and one based on the Bruker E540 console As the spectrometer has developed it has been convenient to have locally designed and constructed modules for all functions of the spectrometer so that components can be changed to test hypotheses about performance When an aspect is well developed it becomes convenient to transfer that function to the Bruker hardware and software system For example at this stage we routinely ac quire FIDs with the SpecJet hardware and Xepr soft ware The magnetic field is controlled by the Bru
33. a tor 356BC Oak Technology Sunnyvale CA The frequency counter was designed and built locally It consists of a prescaler circuit item 41 in Fig 5 and a counter card and a display card Detailed schematic diagrams of the prescaler counter and display cards are available The prescaler divides the source RF by a factor of 256 and provides a transistor transistor logic TTL level signal to the counter card The prescaler also divides the input frequency by 1024 for use in the Bruker frequency counter The counter in the bridge is a gated counter design providing an updated display every 512 ms The measurement gate is 256 ms long The large format LED display ele ments visible on the front panel of the bridge are mounted on the display card Using other prescaler circuits we have adapted this counter circuit to other spectrometers in our laboratory including X band We have also supplied this counter as a stand alone unit to other laboratories Resonator Tuning System Resonator tuning is crit ical for spectrometer performance This is especially true in the case of the CLR in which the frequencies of two resonators must be matched In pulse mode where overcoupling may be desired it is important that the degree of overcoupling and the Q be known It is also important that any tuning mechanism used be such that the resonator and its coupling mechanism stay in the tuned condition when the system is switched from TUNE to OPERATE It
34. age U14 pot R27 because any DC offset will shift the center field The linear sweep ramp is also used in resonator tuning modes to sweep the frequency of the RF source Another use of the linear ramp in our system is to drive the x axis of the x y oscilloscope display This drive is provided through switch U6 Fig 17 Another part of the ramp system is the z drive retrace blanking to the x y oscillo scope This signal is derived from the up down flip flop U3A and sent to the z axis of the scope through connector J3 Field Control in the DU System Center field con trol The center field is set by the user at the PC This becomes a 12 bit value defining a center field over the range of 50 to 150 G Stages U26B and U27B Fig 21 comprise a 12 bit D A converter that pro duces a voltage analog of 0 to 10 V 50 to 150 G scale 10 G V Stage 27C offsets and gain adjusts this signal so that the scale is now 15 G V This stage also sums in contributions from the slow scan system see below and a center field front panel adjustment pot scaled by stage U55D The front panel control can move the center field 10 G Stage U27D adds a microprocessor controllable offset that can be used to compensate for any offset inherent in the magnet power supply The scale factor at the output of U27D is 20 G V with nominally zero offset The signal now goes to Fig 25 where stage U51B changes the scale to 100 G V This is the scale required by the HP p
35. averager These circuits are locally designed and constructed on small printed circuit cards 10 0 cm X 5 46 cm with SMA connectors Pulse Auxiliary Unit Figure 13 shows the pulse auxiliary unit Part num bers and manufacturers for the circled reference num 76 QUINE ET AL FROM VIDEO AMP FIGURE 14 s1 fs R1 49 9 OHMS SELECT 1MHZ 12V SELECT 5MHZ 12V 2 R4 Ja 49 9 OHMS TO BOXCAR v2 V F zor cuc407 TO SCOPE ed a5 49 9 OHMS 1UF LYN 7 3 teas tcai bca L crs cia az zs 1K ax ha 2 T luF 7 10F 7 330F t T 1uF ST 330F 1N4148 G REFLECTED PULSE 2 R2 a 49 9 OHMS INCIDENT PULSE 3 R3 SELECT INCIDENT PULSE N ak 49 9 OHMS 7 l e a szuzct RerzLecTED purse m L CXL Cx TUNE MODE RI R10 LUE DUE 3 3K 1k VED vse 5 om 74AC00 Tex aS 12 T4AC00 FT iur 13 zl 2 8 238 L 10 j R8 S C ON OFF 5 C ON GND ax G Tox Figure 15 Schematic diagram of video distribution circuit card bers are detailed in Table 1 This unit contains most of the components of the power arm of the pulse system except for the power amplifier itself Phase
36. d Smith Chart amplifiers The resonator tuning display and the spectral display are on an x y driven oscilloscope The x axis is driven either by the field sweep ramp spectral display by 82 QUINE ET AL 15V 14 VDD TP23 D7 T DBT VREFA T DIGITAL A 4095 D DBS Say DIGITAL B D5 DBS VREFB R81 tice s0 100 4095 D4 DB4 5K D3 z DB3 RFBA FIELD CENTER oy p2 ps2 pall f 106 R82 T 1Ur 1V 106 D1 DB1 outa A v 34 8K 4v G po pB0 AGN DAI p 15V oE csi cs AD7537 LAB1 16 a DUAL D A ceo c65 LABO 15 x0 RFBB J OGu 3 33V c66 19 oP 1ur 1LUF ov 506 ip WRN 38 WR RBS hipaa Res 5V CLR TPR24 4 99K 0666 6 Ree x asa v TPR25 TE26 u27 LP347 uarn mee R87 LP347 206 7 7 50K TOOK o5v G a 13 oveoc 10v 2006 WA a nn li i 0 0 10v 12 n A 50 TO 1506 J TO PIGURE 25 Laa v vos sys 1 4280 RE6 RI4 NOM 7 65K R86 4 99K R94 4 cw Bois a CAL SCAN RANGE 51023 79 9 77 G V 41sv 10 0K SCAN RANGE 1006 10 23V
37. e On Isolation circles Off Off 14 Pulse CLR Operate Tune 1 Off Off Reflected Pulse Off 15 Pulse CLR Operate CLR Operate Off Off Incident Pulse Off or TD EPR Abbreviations SC Smith Chart Dots fixed not swept frequency display Circ via circulator dir coupler via directional coupler TD time domain 50 mW incident power In both cases the power at the resonator is adjustable over a 70 dB range in dB steps using the attenuator 3 4 front panel control The reference arm of the bridge is used for both CW and 150 300 MHz EPR BRIDGE TUNE STANDBY 7 OPERATE are pulse operation Amplifier 52 provides sufficient power to drive a long delay line in the reference arm The delay line is 22 ft of RG 58U coaxial cable that is external to the bridge box The delay line is neces o CLR AFC PREAMP a OFF SMITH CHART XTAL 2 OFF SINGLE VARACTOR PHASE BIAS MODULATION POWER AMPLIFIER AFC h MODULATION GAIN EXTERNAL 2 LEVEL HIG PONER g WED 4 Pe ae een Low a INTERNAL 1 OFF Sa ER E XTAL 2 YARACTOR REFLECTED POWER XA VOLTAGES DAH wo L HSV 20V 15v O e 0 043 20V zy LIKIT ADDITIONAL ATTENUATION PREAMP PHASE SHFT Ji aN FREQUENCY MHz E ad none CO CAN roy cain zie O 255 882 cw REFERENCE ARM PHASE RESONATOR Figure 9 PULSE CLR OPERATE CONNECTION NE Front panel of DU constructed VHF bridge TUNE fI TUNE 4
38. e other functions of the spectrometer it is more convenient to use a crystal detector as de scribed below for AFC and for finding the resonant frequency for an overcoupled resonator in pulse mode These considerations led to different types of detectors at specific locations in the bridge One can envision that as analog to digital convert ers with requisite speed and resolution become avail able and computer speed increases for postprocessing of EPR signals it may be advantageous to fully dig itize CW and time domain signals at the 250 MHz RF For example in a recent paper from the Hyde laboratory analog to digital conversion of an inter mediate frequency signal was performed at 187 5 MHz in an X band pulsed spectrometer 50 With available technology we use mixers or crystals to convert 250 MHz RF to lower frequencies rapidly changing direct current and then digitize the FID 62 Table QUINE ET AL 1 Components for Optimum Performance Near 250 MHz 1 Fluke signal generator model 6080A 2 13 20 69 20 dB directional coupler Merrimac CRM 20 500 3 4 1 69 dB step attenuator Weinschel 3010 100 5 Amplifier 1 W NF 8 dB typ gain 13 dB min 15 dB typ Miteq AMP 1389 7880 6 Circulator 256 5 MHz UTE CT 1503 0 7 10 3 pole switch Dow Key 435 5208 8 Low noise amplifier 28 4 dB gain NF 1 2 dB Miteq AM 2A 000110 9 Low noise amplifier gain 15 5 dB NF 2 5 dB Miteq AU 1A 0
39. ed to blank the detector to prevent overload of the detector and video amplifiers due to the resonator ring down When the CLR is used with active Q 74 QUINE ET AL PULSE FORMING CONTROL FROM PTU eo REAR PANEL BEC 180 CONTROL Oo REAR PANEL anc o 90 CONTROL J REAR PANEL ot 5V 5v 5V penay O 63 i Ll STEP VARIABLE DUTENE NE PULSAR QUADRAPHASE s o ATTENUATOR ATTENUATOR POWER AMPLIFIER MODULATOR 4 EARR PERR o p0 an 0720 aB BNC a s PINTER WEINSCHEL WEINSCHEL t NL MO B2 412 50 OHM 116A 60 66 910 20 11 FRONT PANEL FRONT PANEL BIDE el L MINICIRCUITS SHP 100 e MINICIRCUITS GaAs SWITCH ZASWA 2 S50DR 15 PIN MALE ELDOR CONTROL REAR PANEL FROM PTU sye i as 5v R F FROM BRIDGE L 5V 5v REAR PANEL SMA bl 2 15V Mg E R F i FROM ELDOR SOURCE REAR PANEL SMA MINICIRCUITS Z SWA 2 50DR i WERLATONE C6117 3 3 FROM POWER AMPLIFIER TO BRIDGE CROSSED DIODE CHANNEL MICROWAVE N TYPE Q NOISE BLANKING 1 AU369 2 7 SMA REAR PANEL EC 17012 ISOLATOR REAR PANEL LEFT SIDE ai 50 dB L INCIDENT PULSE MONITOR TO OSCILLOSCOPE LN 4 REAR PANEL ae ALAN INDUSTRIES an 50D 1 Figure 13 Block diagram of pulse auxiliary box spoiling we have found that the detector blanking function is not necessary T
40. equency by a lock in amplifier and digitized by a locally built box that utilizes the Stanford Research model SR245 13 bit digitizer The field modulation system utilizes a res onant drive to the modulation coils with a switch selection of two different modulation frequencies available 19 and 45 kHz The modulation amplifier and linear sweep amplifiers that drive the modulation coils are described below The lock in amplifier used at 19 kHz is the Stanford Research model SR510 A Stanford Research model SR844 is used at 45 kHz Commercial lock in amplifiers were used because they have the needed functionality and are cost effec tive relative to building these functions into the bridge Although 100 kHz magnetic field modulation is commonly used in commercial EPR spectrometers a current focus on narrow line spectra makes it nec essary to chose lower modulation frequencies whose sidebands are within the linewidths of the experimen tal spectra Bruker CW mode Figure 2 shows the system com ponents used in mode 2 the Bruker CW mode In this mode the Bruker E540 console is used to control the field supply the field modulation and do the data acquisition In this mode no low frequency oscillo scope spectral display is available The E540 controls the 81 cm magnet and magnet power supply Kepco model ATE 36 30M through an interface circuit locally designed and built described separately be low and the Hall sensor described elsewhere i
41. for imaging large biological specimens Meas Sci Technol 1994 5 793 796 9 Symons MCR Whole body electron spin resonance imaging spectrometer In Ohya Nishiguchi H Packer L editors Bioradicals detected by ESR spectroscopy Basel Birkhauser Verlag 1995 p 93 102 10 Bourg J Krishna MC Mitchell JB Tschudin RG Po hida TJ Friauf WS Smith PD Metcalfe J Harrington F Subramanian S Radiofrequency FT EPR spectros copy and imaging J Magn Reson B 1993 102 112 115 11 Pohida TJ Fredrickson HA Tschudin RG Fessler JF Krishna MC Bourg J Harrington F Subramanian S High speed digitizer averager data acquisition system for Fourier transform electron paramagnetic resonance spectroscopy Rev Sci Instrum 1994 65 2500 2504 12 Murugesan R Cook JA Devasahayam N Afeworki M Subramanian S Tschudin R Larsen JA Mitchell JB Russo A Krishna MC In vivo imaging of a stable paramagnetic probe by pulsed radiofrquency electron paramagnetic resonance spectroscopy Magn Reson Med 1997 38 409 414 13 Murugesan R Afeworki M Cook JA Devasahayam N Tschudin R Mitchell JB Subramanian S Krishna MC 90 14 15 16 17 18 19 20 21 22 23 24 25 26 27 28 QUINE ET AL A broadband pulsed radio frequency electron paramag netic resonance spectrometer for biological applica tions Rev Sci Instrum 1998 69 1869 1876 Rubinson KA Cook JA Mitchell JB Murugesan R Krish
42. gi LP347 10K a 3 R103 20 0K 83 RESET TO BRIDGE RESSW i a7 OM MW 1 P m 50 R152 LOK MWH2 e126 cris 1UF 1N4148 US8A A T4ACTA PR Sew e a9 v C7711 15V e 1UF v34 R105 R106 10 0K CRE 7oxnza 4 SPI R L GND GND e 7 DBM GND GND X1 x2 PROM FIGURE 187 5 3 4 R104 20K ao c78 U33B LF347 o1naias METER ZERO e u56 LM7812 OUTI GND 412V 1UF R102 LOOK u33D R101 LF347 CR7 1N4148 c1is T 330uF e115 T 1uF 200K 13 TP33 15V u57 LM791 GND 2 OUT r 12v 12 IH L 123 1UF 3 c124 l 125 1uF 7 33UF Figure 22 Lock In Amplifier We currently use two different lock in amplifiers depending on the field modulation frequency At 19 kHz we use an analog type Stanford Research model SR510 and at 45 kHz we use a digital lock in Stanford Research model SR844 The SR844 does not operate below 25 kHz These units provide phase sensitive detection at the field mod ulation frequency adjustable gain and final signal filtering before being digitized We digitize the output of either lock
43. h plans to eventu ally use the Bruker PatternJet timing system The spectrometer described has achieved the goal of facilitating testing of both LGR and CLR in CW and pulse operation with two magnets and multiple power supply combinations The FID signal ampli tude agrees with prediction based on the properties of the resonator and the measured overall gain of the detection system and agrees with predictions as a function of RF microwave frequency 49 Further development is planned of enhancements in resonators to decrease dead time and of improve ments in RF power amplifiers to increase pulse power and decrease dead time It is also planned to place a CLR and first stage amplifier in a cryostat and cool them to liquid He temperatures for echo envelope modulation studies of proteins If only one of the functions built into this bridge is needed e g CW with reflection resonator or pulse with CLR the bridge could be simplified consider ably ACKNOWLEDGMENTS A generous gift of trityl radical from Nycomed Inno vations AB to Professor Howard J Halpern Chicago facilitated this research Initial construction of the prototype CW bridge and the 40 cm diameter magnet was funded by National Science Foundation STTR Grant DMI 9523205 to Omni Engineering and the University of Denver in 1995 Design and construc tion of the resonators used in Denver was supported by National Institutes of Health Grants RR12183 and GM57577 G A R The
44. he detector blanking switch is controlled by the PTU and its time position can be set by a combination of software settings in the PC and an analog delay knob on the front panel of the PTU In pulse mode two different detection systems are available by bridge front panel selection In NOR MAL DBM 14 is the detector and all of the available signal is directed to this detector Normally this sin gle channel detection system is used in the DU pulse mode in which data are collected in the boxcar aver ager In the Q or quadrature mode power splitter 89 Fig 8 divides the signal between two DBMs 90 and 91 These mixers operate in quadrature from a refer ence signal that is divided by 90 hybrid splitter 92 This produces detected signals J and Q which are amplified by video amplifiers and routed to the Bruker SpecJet dual channel digitizer Bruker pulse mode CW Low Noise Amplifier Amplifier 21 Fig 8 is used to amplify the CW EPR signal before it goes to the lock in This amplifier is of local design and construction and is of similar design to 70 kHz AFC preamplifier 24 These amplifiers are built on small circuit cards 6 9 cm X 4 3 cm and packaged in small shielded boxes The low frequency response of this amplifier has been extended below 1 kHz so that low frequency modulation can be used The am plifier is designed with low noise transistors type MPS4356 to achieve the best possible noise perfor mance however this amplif
45. he lower edge of Fig 21 is dual digital to analog D A converter U28 Stage U28B converts a 12 bit number that represents the mantissa of the desired modulation amplitude set by the user at the PC into an analog voltage in the range of 0 to 5V This voltage indicated as CMD in the figure is then sent to stage US5A lower edge of Fig 20 Stage PULSED AND CONTINUOUS WAVE 250 MHz EPR SPECTROMETER 79 SAWTOOTH CAL R24 9 53K f 225 in 2K cw ure R23 LF347 FROM FIGURE 17 20 0K r NW FROM BRIDGE LEMO 65Gpp 5 6Vpp 7 TO POWER AMP RFBB l 15V 2 4 15V TO LOCK IN J18 FROM SR 844 OUTPUT 70RHZB TO FIGURE 22 TP7 Jigs TO SR 245 5 z R31 JAAco4 BRIDGE CONNECTOR S 10K 10F 1 Poot SCMODE apf 9 eee R35 5ORHz 15V SWEEP RAMP ake 402K Hee 3 200 r 5V A us 10 a js vs a Js 4 20V TETT v vee a R36 32 rE MODE SWEEP 2 pe419 R34 a 5 4 99K 13 Y DRIVE TO SCOPE 12 4 4 gt SMITH CHART B 8 o 12 Loj i SCMODE s S CHART A SoA a v GND Y D k 7 E TUNE MODE 7 13 piney E 3 FE n R32 mesy cro aon Lis ale 14t 4 16K 15V eri
46. ier will make a significant contribution to the noise floor only if no low noise RF preamplifier is used The nominal gain of amplifier 21 is 40 dB 100X Video Amplifiers Video amplifiers 93 and 62 Fig 8 are used to amplify the detected signals in pulse mode before they are sent on to either the boxcar averager or the SpecJet digitizer and a monitoring oscilloscope The amplifiers are of local design and construction These amplifiers are gain adjustable from the bridge front panel in four steps of 34 40 46 and 54 dB 50 100 200 and 500X They also have DC offset controls on the front panel We have experimented with both AC and DC coupling for these amplifiers and find there are tradeoffs and a case can be made for either coupling method We are currently using AC coupling with the high pass corner at 10 Hz and PULSED AND CONTINUOUS WAVE 250 MHz EPR SPECTROMETER 75 V IN 15V 15V L2 El 1000H U5 LM78050P IN wp Oop 5Y c22 court ore Lcis giy cis ae des zg m T 10F T 10r T 330F T io LE spt 330P RIA F 499 5v o23 RP1 10K GAIN CONTROLS anag 4 GROUND TO SELECT 13 fa 15pf L3 v vec R9 1000H us 200 s00 A il al 3a aj 5 6 4PS432 WA war GRY L4 5v L as EG 1000H 1p eet Q ANNA
47. in using a Stanford Research model SR245 digitizer but we could also obtain digital information directly from the SR844 We have not implemented this feature Modulation Power Amplifier The modulation power amplifier Fig 27 is based on an Apex model PAOS power op amp This am plifier has a gain bandwidth product of 400 kHz when driving a 4Q load and can drive loads of up to 10 A with up to 40 V We used the large heat sink and printed circuit board supplied by Apex EK04 We set the gain at 10X which is sufficient to drive the 7 inch 17 8 cm modulation coils when reso nated with a series capacitor Figure 27 shows the resonating capacitor selection switch which is on the front panel of the unit There is also a current reversing switch that can compensate for mounting the modulation coils in the main magnet in either of two possible orientations Inductor L1 is locally constructed It is wound with no 12 AWG magnet wire on a laminated iron core An air gap was inserted to prevent saturation from the nearly DC current The inductance is 31 mH This inductor isolates the sinusoidal modulation system from the low frequency linear sweep system Linear Sweep Power Amplifier The linear sweep power amplifier Fig 28 is a cur rent mode amplifier of local design and construction Current is sensed by resistor R9 and U1B produces a voltage proportional to current of 0 5 V A This is the feedback to the main driver stage U1A so
48. ineering 15 59 91 2002 KEY WORDS EPR imaging spectrometer design pulsed EPR INTRODUCTION Electron paramagnetic resonance EPR in the tens to hundreds of megahertz range has been performed in many laboratories beginning with the discovery of EPR The majority of EPR spectra are obtained at X band 9 GHz There are many incentives for obtaining EPR spectra at frequencies below X band Resonator structures can be made to accommodate larger samples e g mice at lower frequencies The tradeoffs between sample volume resonator filling factor and microwave loss factor can be optimized at frequencies below X band for some lossy samples Received 10 October 2001 revised 11 November 2001 accepted 12 November 2001 Correspondence to Gareth R Eaton E mail geaton du edu Concepts in Magnetic Resonance Magnetic Resonance Engineer ing Vol 15 1 59 91 2002 2002 Wiley Periodicals Inc including biological aqueous samples The tradeoffs between g anisotropy and a anisotropy can result in the minimum linewidth and hence the best hyperfine resolution occurring at frequencies below X band Some electron spin relaxation mechanisms are mag netic field dependent 2 Measurements of relaxation times at a wide range of magnetic fields are needed to provide a test of mechanism Swartz and Halpern 3 recently reviewed in vivo EPR comprehensively cov ering the low frequency literature Greenslade et al 4 c
49. is anticipated that a wide variety of resonators including reflection and crossed loop resonators will be tested with this spectrometer Experience with one CLR is described in an accompanying paper For these reasons we have built into the bridge an extensive set of resonator tuning facilities Smith Chart Display The bridge incorporates a Smith Chart coupling dis play which we have found to be generally more useful and easier to interpret than the customary dip in the display of the reflected power vs frequency 57 The reflected signal is obtained through circula tor 6 Fig 5 in modes in which the circulator is required in OPERATE i e all LGR modes and the CLR CW mode In modes in which the circulator is not used in operation i e pulse CLR modes direc tional couplers 49 and 74 Figs 6 and 7 are used to obtain reflected signals for resonator tuning Transfer switches 33 and 34 switch to the appropriate signal paths for tuning the various resonator configurations When the Smith Chart display is in use coaxial switch 19 redirects the reference arm power to Smith Chart mixers 28 and 29 These mixers operate in quadrature from 90 hybrid coupler 27 and in phase power divider 30 The intermediate frequency IF port of each of these mixers is directed to amplifiers in the microprocessor interface box see details below and ultimately to the x y display oscilloscope The Smith Chart can optionally be operated in the c
50. ker ERO32T field control system which uses a Hall probe The device developed to use this to control the current in the air core magnet is described in an accompanying paper 59 Numerous CW and pulsed EPR spectra have been run to test the functions of the spectrometer and challenge its performance Some FIDs and spin echo measurements of relaxation times obtained with the spectrometer are presented in Ref 40 The sample was a 0 2 mM aqueous solution of the deuterated symmetric trityl and the trityl called OX 31 both synthesized by Nycomed 53 The samples contained 4 6 X 10 spins in a 10 mm outside diameter 0 d NMR tube A larger sample could have been used since the aqueous sample causes only a small reduc tion of resonator Q at 250 MHz The echo signal in 86 QUINE ET AL FROM FIGURE 21 Rial 4 99K A 20G O5V G ova0c 10V 2000 TP43 c10s 15V 4 Ea si gt 1ur laa asy V 15sy Toia R143 LM319 4 7K 1oomMVv oc R149 EE 10k 12 MODOVTMP 2 v cnND e 3 HILOVOLT oefe 3 4 RESCAP 15V Lo 5 R144 LE 470K EA n eo a 4 R145 st S1ox 5V i TP44 F U54B kiat i LM319 4 7K R148 10 i 20K sf gt ranere cw GND T TP42 15sv zt R147 i 470K 7 AWA Fig 8 of Ref 40 obtained by accumulating 99 328 echoes in 13 4 s using the Bruker SpecJet digitizer had S N 396 The pulse
51. mode the spec 64 QUINE ET AL FLUKE i BRIDGE PHASE TUNE METER SIGNAL CHANNEL RF GENERATOR MODULATION OILS MAIN MAGNET corns DISPLAY FOR TUNE BRUKER E540 CONSOLE MODULATION AMPLIFIER g FIELD CONTROL HALL PROBE SENSOR MAGNET POWER INTERFACE SUPPLY CIRCUIT Figure 2 Block diagram of Bruker CW mode trometer is normally operated as a conventional CW EPR spectrometer with field modulation The modu lation coils are also used to provide a low frequency 10 Hz linear field sweep for the purpose of pro viding an oscilloscope spectral display The field modulation and magnet are controlled from a PC using locally written software The modulation sys tem the 10 Hz sweep system the control of the magnet power supplies and various tuning and dis play functions are performed by a locally designed and constructed microprocessor controlled interface unit described below The field modulation fre quency is supplied from an HP model 3310B function generator Most of the bridge functions are controlled manually from the bridge front panel The PC also controls the data acquisition and performs the postac quisition data workup Spectral information is demod ulated at the field modulation fr
52. n the bandwidth PULSED AND CONTINUOUS WAVE 250 MHz EPR SPECTROMETER 61 of the circulator Isolators which in general are used to absorb the power from a mismatched load and which help reduce the impact of switching transients on other active devices are circulators with one port terminated in a matched load Consequently isolators have the same bandwidth limitations as do circulators Greater flexibility in choice of frequency is desirable and is an advantage in the design of some prior spectrometers There is a general tradeoff between bandwidth and other performance features of many components used in the bridge We selected components for optimum performance near 250 MHz Of the components listed in Table 1 a large fraction have bandwidths narrow enough to prevent increasing or decreasing the oper ating frequency by as much as a factor of 2 A dominant feature of the design philosophy for this spectrometer was maximum flexibility to test alternative approaches to both CW and pulsed EPR and to test various magnets power supplies RF sources RF amplifiers resonators etc This require ment resulted in considerable complexity that was only slightly decreased by modular construction It became evident that the incident RF and EPR signal paths had to be switched in so many combinations that the losses would be unacceptable Consequently we adopted as a central design feature the use of small plug in circuit cards containing microstri
53. n this issue Control of the 40 cm magnet with the E540 has not yet been implemented The Bruker system also supplies the lock in function through its normal signal channel The DU bridge and resonator configurations i POWER PULSED AND CONTINUOUS WAVE 250 MHz EPR SPECTROMETER AMPLIFIER PULSE AUXILIARY UNIT FLUKE RF GENERATOR MAIN MAGNE corns BRIDGE BOXCAR AVERAGER DI SR245 GITIZER NJ PREAMPLIFIER X X DISPLAY FOR TUNE PROGRAMMABLE TIMING UNIT PENTIUM PE 65 OR CLR Q SPOIL MICROPROCESSOR INTERFACE UNIT MAGNET POWER SUPPLY Figure 3 Block diagram of DU pulse mode available are the same as in mode 1 The modulation coil driver system in the Bruker console was designed for much smaller lower inductance modulation coils than we use here the 7 inch 17 8 cm coils described below Therefore only modest modulation ampli tudes are possible even when the nominal modulation amplitude is set to very high values There are reso nator capacitor selections that approximately resonate the 7 inch coils up to about 50 kHz and down to about 15 kHz but at some frequencies the Bruker capacitor selections are too widely spaced to obtain perfect resonance We also
54. na MC Subramanian S FT EPR with a nonreso nant probe use of a truncated coaxial line J Magn Reson 1998 132 255 259 Subramanian S Murugesan R Devasahayam N Cook JA Afeworki M Pohida T Tschudin RG Mitchell JB Krishna MC High speed data acquisition system and receiver configurations for time domain radiofrequency electron paramagnetic resonance spectroscopy and im aging J Magn Reson 1999 137 379 388 Afeworki M Gooitzen MvD Devasahayam N Murug esan R Cook J Coffin D A Larsen JH Mitchell JB Subramanian S Krishna MC Three dimensional whole body imaging of spin probes in mice by time domain radiofrequency electron paramagnetic resonance Magn Reson Med 2000 43 375 382 Devasahayam N Subramanian S Murugesan R Cook JA Afeworki M Tschudin RG Mitchell JB Krishna MC Parallel coil resonators for time domain radiofre quency electron paramagnetic resonance imaging of biological objects J Magn Reson 2000 142 168 176 Blume RJ Electron spin relaxation times in sodium ammonia solutions Phys Rev 1958 109 1867 1873 Sachs G St cklein W Bail B Dormann E Schwoerer M Electron spin relaxation of new organic conductors fluoranthenyl radical cation salts Chem Phys Lett 1982 89 179 182 Sachs G Dormann E Low field pulsed electron spin resonance in organic conductors Bruker Report 1984 p 30 Callaghan PT Coy A Dormann E Ruf R Kaplan N Pulsed gradient spin echo ESR J Magn Reson A 1994 111 127 131 Co
55. nce of nitroxide spin labels Science 1985 227 517 519 Fujii H Berliner LJ One and two dimensional EPR imaging studies on phantoms and plant specimens Magn Reson Med 1985 2 275 282 Koscielniak J Berliner LJ Dual diode detector for homodyne EPR microwave bridges Rev Sci Instrum 1994 65 2227 2230 Nilges MJ Walczak T Swartz HM 1 GHz in vivo ESR spectrometer operating with a surface probe Phys Med 1989 2 4 195 201 Froncisz W Hyde JS The loop gap resonator a new microwave lumped circuit ESR sample structure J Magn Reson 1982 47 515 521 Hyde JS Froncisz W Loop gap resonators In Hoff AJ editor Advanced EPR applications in biology and bio chemistry Amsterdam Elsevier 1989 p 277 306 Rinard GA Quine RW Ghim BT Eaton SS Eaton GR Easily tunable crossed loop bimodal EPR resonator J Magn Reson A 1996 122 50 57 Rinard GA Quine RW Ghim BT Eaton SS Eaton GR Dispersion and superheterodyne EPR using a bimodal resonator J Magn Reson A 1996 122 58 63 Rinard GA Crossed loop resonator structure for spec troscopy US Patent 5 739 690 1998 Rinard GA Quine RW Eaton GR An L band crossed loop bimodal resonator J Magn Reson 2000 144 85 88 Rinard GA Quine RW Eaton GR Eaton SS 250 MHz crossed loop resonator for pulsed electron paramag netic resonance Concepts Magn Reson 2002 15 49 58 Sotgiu A Gualtieri G Momo F Indovina PL ESR imaging an overview Phys Med 1988 3 4 149
56. netic field gradients in low frequency EPR imaging systems 4 45 It is not necessary to build a spectrometer if the goal is spectroscopy and imaging of small objects at L band since a fully functional spectrometer that takes advantage of some of the software developed for magnetic resonance imaging is available from Bruker BioSpin EPR Division The Elexsys E540 L band imaging system performs two and three dimensional spatial imaging and spectral spatial imaging The res onator has 34 mm free access diameter The spectrometer described in this paper differs from prior spectrometers in the design of the bridge magnet resonator and pulse timing system This spectrometer has been designed for maximum flexi bility for testing alternative components and will be used in the future to seek improved performance of pulsed 250 MHz EPR In addition the spectrometer is designed with the intent that as it evolves it will be compatible with commercial spectrometer compo nents to facilitate transfer of technology to other lab oratories MATERIALS AND METHODS The 250 MHz spectrometer builds on prior EPR spec trometer design experience in our laboratory e g 46 49 The spectrometer is designed from an EPR perspective and differs from typical NMR spectrom eters in spite of operating at a frequency common in NMR Thus for example the bridge is built around a circulator as is common in microwave frequency EPR spectrometers Although the
57. ntrol automatic frequency control AFC and the strip line resonator The bridge design uses high dynamic range double balanced mixers to accommo date animal induced changes in reflected power The CW L band spectrometer in the Berliner lab oratory has been described 29 32 There have been many reports of low frequency EPR spectroscopy from the Swartz laboratory whose 1 GHz spectrom eter was briefly described in Nilges et al 33 Sy mons 9 has given a brief description of a 250 MHz spectrometer intended for whole body human imag ing The overall design is similar to that of a smaller 300 MHz system 7 One of the resonators is a 45 cm diameter three loop two gap loop gap resonator LGR the two outer loops are 10 cm The Q of 300 was reduced to 100 upon inserting a human head The Symons 300 MHz spectrometer was de signed for samples up to about 5 cm radius An enabling technology for low frequency EPR is that of lumped circuit microwave resonators that per mit a higher filling factor and higher B per watt than cavity resonators at these frequencies 34 Rectangu lar cavity resonators or cylindrical cavity resonators would be extremely large at these low RFs see the review by Hyde and Froncisz 35 The crossed loop resonator 36 39 used for pulsed EPR in our spec trometer is described in an accompanying paper 40 Sotgiu and coworkers incorporated reentrant resona tors and multipole magnets which can produce mag
58. on ventional single trace mode or in a biphase chopped mode The chopped mode is useful because it elimi nates the need for a calibrated graticle on the display In the chopped mode there are two traces 180 apart Critical coupling is indicated when the two traces are tangent Overlapping traces indicate overcoupling and underlapping traces indicate undercoupling The chopping signal is developed by an oscillator on cir cuit card 26 The chopper frequency is adjusted to approximately 5 kHz The exact frequency is chosen so that the beats between the chopper frequency and the frequency modulation FM sweep produce a smooth display The chopper frequency is logically combined into the path of the 180 bit of the discrete phase and sent out to the phase shifter by a 500 driver Resonator O Dip Displays Although we normally depend on the Smith Chart for resonator tuning we occasionally find that a traditional resonator dip display is useful The bridge produces a PULSED AND CONTINUOUS WAVE 250 MHz EPR SPECTROMETER 71 dip display detected by crystal 79 LGR or crystal 80 CLR Fig 5 This detected signal is amplified by the same amplifier 24 used for the AFC preamplifier and routed through circuit card 26 where it is ampli fied further and provided with direct current DC offset control From here it is routed to the micropro cessor control box where it is ultimately sent to the y axis of the x y oscilloscope display Similarly a
59. onents and careful posi tioning of the magnet relative to the bridge minimized the baseline slope For pulsed EPR it is necessary to amplify the source RF One approach is to use a CW amplifier pulse the input and use extensive signal averaging to reduce the noise that is output by the RF amplifier during the signal detection period For our initial testing we have done this using a Dressler 75 W CW class A amplifier which at 250 MHz delivers 20 W to the resonator FIDs have been observed for Nycomed trityl and lithium phthalocyanine radicals and spin echoes have been observed for trityl and irradiated SiO We have used several different style resonators at 250 MHz A small five turn coil sized for 4 mm o d tubes similar to an NMR resonator was used for mapping the magnetic field with small samples A LGR that holds a 1 inch 25 mm o d sample tube analogous to the LGRs we have used previously at higher frequencies 49 has been used for CW spectra with magnetic field modulation A CLR analogous to those we used at S band and L band but large enough to hold a 1 inch o d sample tube has been used for pulsed EPR where the isolation is important 40 This spectrometer incorporates two air core mag nets One has four coils of the same diameter with a cylindrical sample space of 40 cm diameter The other magnet described in Ref 60 is about twice the diameter and achieves a 15 cm diameter homoge neous volume This magnet al
60. ower supply which drives the 40 cm magnet see below Stage U51C rescales to 142 G V which is the scale required by the Kepco ATE supply that drives the 81 cm magnet Switch U52 selects between the HP and the Kepco scale factors This selection comes ulti mately from the user at the PC Switch U53 turns the center field on or off again a selection of the user at the PC The final output at J10 is cabled to either the HP or the Kepco power supply depending on which magnet is in use Slow field scan system Stage U25A Fig 21 inverts the scan voltage arriving through connector J6 from the SR245 13 bit D A converter This was de signed to be compatible with our previously designed software that controls our other spectrometers through SR245 interfaces The slow scan software in the PC controls the SR245 over a serial communications port which then provides a 10 24 to 10 23 voltage scan over the scan range selected Stage U28A and U29A PULSED AND CONTINUOUS WAVE 250 MHz EPR SPECTROMETER 81 R59 R60 10 0K 30 1K n e RES cw 4 02K AM 5 VAN R68 LIA REF 1K R67 ow 2 0K WWA 5VAN ivpp SINE 592V c40 Pd nas Lt 5K ies L 5VAN een asor 1 908V TH 10r ozsa ha n can c46 vop TP19 1 TP45 U24A ive F
61. p circuits that redirect the circuit paths for each of the four combinations of LGR crossed loop resonator CLR CW and pulse The circuit cards can be changed quickly and this can be done with power on if the bridge is in standby mode Most components in the bridge are connectorized for ease of replacement or interchange and have SMA connectors Most RF excitation and signal paths in side the bridge use 0 141 inch flexible coaxial cable or semirigid coaxial cable Outside the bridge RG 58 U coaxial cable with BNC connectors is used for most signal paths Paths from the RF source to the bridge from the bridge to the resonator and in the case of the CLR from the resonator to the detection system use s inch diameter coaxial cable type RG 8U with BNC type connectors which has much lower loss than does RG 58U cable General The VHF 250 MHz EPR spectrometer designed and built by the University of Denver DU consists of a number of subsystems and operates in a number of different modes Each of these subsystems will be described The operational modes optionally use ei ther the DU designed data acquisition and control system or the Bruker E540 console or a combination of both There is a choice of two different magnets an 81 cm air core magnet described separately in this issue and a 40 cm air core magnet As described there the magnet was designed to have good homogeneity over a large volume of sample and to provide ea
62. pulsed spectrometer design and development was supported by National Institutes of Health Grant P41 RR12257 H J H and G R E and by Bruker BioSpin EPR Division and the Univer sity of Denver REFERENCES 1 Eaton GR Eaton SS EPR spectrometers at frequencies below X band Biol Magn Reson in press 2 Eaton SS Eaton GR Relaxation times of organic rad icals and transition metal ions In Eaton GR Eaton SS Berliner LJ editors Distance measurements in biolog ical systems by EPR Biol Magn Reson 2000 19 29 154 3 Swartz HM Halpern H EPR studies of living animals and related model systems in vivo EPR In Berliner LJ Spin labeling the next millennium Biol Magn Reson 1998 14 367 404 4 Greenslade DJ Koptyug AV Symons MCR Aspects of low frequency low field electron spin resonance Roy Soc Chem Annu Rep C 1996 92 3 21 5 Halpern HJ Spencer DP van Polen J Bowman MK Nelson AC Dowey EM Teicher BA Imaging radio frequency electron spin resonance spectrometer with high resolution and sensitivity for in vivo measure ments Rev Sci Instrum 1989 60 1040 1050 6 Halpern HJ Bowman MK In Eaton GR Eaton SS Ohno K editors EPR imaging and in vivo EPR Boca Raton FL CRC Press 1991 p 45 63 7 Brivati JA Stevens AD Symons MCR A radiofre quency spectrometer for in vivo imaging J Magn Re son 1991 92 480 489 8 Stevens AD Brivati JA A 250 MHz EPR spectrometer with rapid phase error correction
63. r is amplified by amplifier 5 to the 30 dBm 1 which it is desired to have the lowest noise source W level After losses in switching elements cables power available and which do not require more than FRON FIGURE 5 RI Grom amac e 6 aB cs PRON FIGURE 5 x Cee ae aot to SR 244 or Varian me x100 LENG e I a BNC T al 33 PAN A l Bi a oxone on araamtain NORMAL I amp Q yrn AMP aa scope swrrcrtnc 4 aj ea ana a I CHANNEL P ae PP ee G DBM F 0 SPLITTER R Gmm i 90 HYBRID SPLITTER 4 AN W Q CHANNEL Je ces eee a JP ake se so Figure 8 PULSED AND CONTINUOUS WAVE 250 MHz EPR SPECTROMETER 69 Table 2 Operational Modes of the Bridge Smith Chart Plug In Tune Operate Resonator On Off Pulse Scope AFC Mode Module Switch Mode Switch Switch x y Scope Display Display Operation 1 CW LGR Tune LGR On SC or Dots Circ Off Off 2 CW LGR Tune LGR Off XTAL dip Off Off 3 CW LGR Operate LGR Off EPR Off Locked to LGR 4 CW CLR Tune Tune 1 On SC or Dots Circ Off Off 5 CW CLR Tune Tune 2 On SC or Dots dir coupler Off Off 6 CW CLR Tune CLR operate On Isolation circles Off Off 7 CW CLR Operate CLR operate Off EPR Off Locked to 1 Pulse LGR Tune LGR On SC or Dots Circ Off Off 9 Pulse LGR Tune LGR Off XTAL dip Off Off 10 Pulse LGR Operate LGR Off Off I R or TD Off EPR 11 Pulse CLR Tune Tune 1 On SC or Dots dir coupler Off Off 12 Pulse CLR Tune Tune 2 On SC or Dots dir coupler Off Off 13 Pulse CLR Tune CLR operat
64. re are many ways to detect electron paramagnetic resonance we have cho sen to restrict the design of the spectrometer described here to two fundamental types of measurements CW detection with magnetic field modulation and phase sensitive detection at the modulation frequency and electron spin echo ESE detection These choices were based on the general experience that these meth ods yield the best signal to noise ratio S N and with a view toward applications that include CW and pulsed EPR in vivo imaging electron spin relaxation time measurements and electron spin echo envelope modulation for determination of nuclear electron in teractions in proteins The choice of 250 MHz is based on the considerations of depth of penetration into animals outlined by Halpern and Bowman 6 A circulator efficiently routes the source power to the resonator and the EPR signal from the resonator to the detector Either but not both of these paths can be made low loss with a directional coupler The primary disadvantages of using ferrite circulators and isolators in a very high frequency VHF 30 300 MHz bridge are that they are narrow band and are sensitive to changes in the magnetic field Although octave band width circulators and isolators are available above 1 GHz in the VHF region bandwidths commonly are 10 This requires that the operating frequency be chosen early in the design and hence requires that the resonator be designed to work withi
65. repetition rate was limited FROM FIGURE 19 RX C NOM GAIN 0 7037 NOM R137 2K CALIBRATE ATE R136 A 99K R137 GAIN ADJ FROM 0 5 TO 1 5 L cw NOMINAL FOR 19A e 90G aos ATE 30A 033333 V A usic 2142 G v LP347 0 007037 V G TP38 2 8 R134 10 0K r R138 016 V G 10 0K 100 G V R132 1 24K 15V 5V 15V 5V usa Js MEW MAGNET us2 a s V voc TP40 ATE SEE ABOVE v vec 2 D6419 J10 ATE 2 paai9 Rd PB E i Boon slot Bee AIRCORE v GND v GND H P 7 3 HP aa AY 7 Ja OLD macuET 00 07 15V 101 Y G 15V 7 OLD MAGNET C107 L0000uUF c1o09 c108 T Lo000ur Figure 25 TI 10000UF C112 clli 4700ur ciio T 47000F 33000F 5VAN 5V 15V 15V 20V 20V 113 T 3300UF by the long relaxation time of this species The CLR was overcoupled as described in Ref 40 and the echo signal was amplified by the Berkshire amplifier close cx 1UF uxa Y L OP467 7 2 Be 162K cx gt 1 3 MAC gt Tea ior ge 1N4148 pee 11 L 7 RX 15V ak 51 1K ls ea T 10F uxc OP467 a 10 2 pe TO FIGURE 20 UXB 0P467 Dx RX 7 162K 5 p aaan amma Lex 1N4148 T 4 7UF RX bes 51
66. rior to the amplifier establishes the noise floor of the spectrom eter and hence yields the best S N achievable with a given low noise amplifier We use a cryogenically coolable amplifier model U 250 2 obtained from Berkshire Technologies Oakland CA item 75 in Fig 7 to facilitate future measurements in which the resonator and the low noise amplifier will be cooled to approximately the temperature of the sample to further improve S N 52 This amplifier has 39 dB WHITE TO RELAY DRIVER Figure 12 Schematic of protection limiter circuit gain at 250 MHz To facilitate tuning of the resonator directional coupler 74 is required between the ampli fier and the resonator Low Noise Preamplifiers Amplifiers 8 and 9 Fig 5 are low noise amplifiers Miteq AM 2A 000110 and AU 1A 0150 respectively Low gain amplifier 9 with 15 dB gain can often be used in addition to the Berkshire external amplifier with the CLR This pro vides a total of 54 dB of gain before the detector In some cases the external amplifier alone gives the best results and the NONE preamplifier path is then se lected on the bridge front panel Amplifier 8 with 28 dB of gain is more often used with an LGR where no external amplifier can be used Detectors and Detector Blanking In the CW mode double balanced mixer 14 Fig 8 is the detector Detector blanking switch 70 Fig 5 is not used in CW mode In LGR pulse mode blanking switch 70 is often us
67. ritically reviewed low frequency EPR and dis cussed the relative benefits of various approaches EPR spectrometers at frequencies below X band have been reviewed comprehensively In the fol lowing paragraphs we cite selected publications rel evant to the work reported in this paper Most prior spectrometers have been for continuous wave CW operation The most complete descriptions of EPR spectrometers in the 250 300 MHz range are those of Halpern et al 5 6 Symons and others 7 9 and Krishna and coworkers 10 17 These spectrometers 59 60 QUINE ET AL were constructed for the study of biological speci mens Pulsed EPR spectroscopy at low radiofre quency RF was performed at 17 4 MHz by Blume in 1958 78 and more recently has been performed with nuclear magnetic resonance NMR spectrometers 19 26 and with purpose built spectrometers in the laboratory of Krishna 70 17 and Sotgiu 27 28 The Halpern spectrometer used a strip line type resonator 1 25 cm radius that could hold a small mouse and a hybrid coupler to direct RF to and from the resonator 5 6 It used an air core Helmholtz magnet to generate the magnetic field and splayed these coils to create the gradient in the z direction Gradients along the x and y directions were generated with Anderson type gradient coils This spectrometer is described in considerable detail including the structural and electronic aspects of the automatic cou pling co
68. ront panel These are the manual bridge controls that are not computer controlled The front panel of the bridge is 44 8 cm X 31 1 cm and is silk screened black on a white background The choice of manual controls was a fundamental design specification since flexibility and versatility were a high priority and it was desired to be able to change hardware features without having to also change soft ware RF Source Most RF sources exhibit phase noise that will dominate the noise of an EPR spectrometer at most practical power levels The shape of the phase noise vs frequency plot differs depending on the way the source was built One source or another may be preferable depending on the modulation frequency to be used The better the source the higher the power you can operate without S N being limited by source noise The lower the Q of the resonator is the higher the power at which the source noise begins to domi nate The RF source is a Fluke signal generator model 6080A We also used an HP model 8640B as a source but found that because it was not digitally synthesized the frequency was too unstable for this application The Fluke is stable but does not have the very best phase noise specifications that are available in the latest signal generators In our applications to date this compromise in phase noise performance has been acceptable because the primary focus has been on time domain spectroscopy and on use of the CLR Generall
69. rt of the RF source generator The AC component amplitude AFC mod ulation amplitude is set by a front panel control Stage USC Fig 10 also combines the tuning sweep ramp into the FM signal to sweep the frequency when in TUNE mode The AFC source frequency is sup plied from a locally constructed 70 kHz oscillator based on a 70 kHz fundamental mode crystal OT 11 70 kHz obtained from International Crystal Inc Oklahoma City OK Stages ULOC U10D ULIC 72 QUINE ET AL 22 Lge WM LO N R22 10 0K c21 EE il 1UF 7 5V R17 37a R g 10 EE aa i 13 10K 3 ve vec 2 u2 ADG431 wep nas 373 GAIN CONTROL a 3 2 2 49K 1 HI N E Zi i 15 i v R27 v1 ig 4 99K OP27 rae T 623 2 s 6 8 Ht is 001UF v 4 20V ee sweep foz cs R1 42 2K c p 15v 470p c26 _ 10f7 ua ve 24 mo y m c4 v ce i TPT 39pt oy eH Hers 4 1UF 15V R2 zD DS 100K ANM SIG1 1N4686 1N4148 3 9v Ria Raa TO AFC METER 470 OHM 1 74K EA ZD2 1N4148 1N4686 pa de 228 3 90 1N4148 1N4148 D1 D4 LJ c24 1N4148 1N4148 1UF R4 TPa
70. s These diodes switch fast enough that at 250 MHz they perform a real time single cycle clipping function when the power is above 6 dBm With an input power of 20dBm the insertion loss of the limiter is not measurable When the input power is 0 dBm the insertion loss is 0 6 dB At 20 dBm the insertion loss is 11 dB When the diodes are conducting the limiter becomes a mismatched load that reflects most of the incoming power back to the source where it is absorbed in isolator 57 Fig 13 pulse mode or isolator 83 Fig 5 CW mode The limiter has been tested to survive and operate properly with 400 W pulses of 1 ms duration and 1 duty cycle and with 1 W of CW power Limiter 98 also contains a DC sense circuit that detects the presence of either a pulse configura tion module or a CW configuration module The lim iter is constructed on a small printed circuit card 3 81 cm X 3 75 cm of local design and housed in a small box with SMA connectors External Low Noise Preamplifier The isolation of the CLR replaces the circulator function in a conven tional EPR spectrometer designed for use with a re flection resonator 36 40 The CLR configuration shown in Fig 7 allows the possibility of having the first low noise amplifier in the signal path to be very near the resonator This location for the low noise RF R67 PULSED AND CONTINUOUS WAVE 250 R66 100K MHz EPR SPECTROMETER 73 70 KHz AMPLITUDE TPIS R73 4 99
71. s Magn Reson 2002 15 59 62 Rinard GA Quine RW Eaton GR Eaton SS Barth ED Pelizzari CA Halpern HH Magnet and gradient coil system for low field EPR imaging Concepts Magn Reson 2002 15 63 70 Eaton GR Quine RW Comparison of four digitizers for time domain EPR Appl Spectros 2000 54 1543 1545
72. se saturation recovery pulse ELDOR and free induction decay electron paramagnetic resonance detection using time locked subsampling Rev Sci Instrum 2001 72 1837 1842 Rinard GA Quine RW Eaton SS Eaton GR Froncisz W Relative benefits of overcoupled resonators vs in herently low Q resonators for pulsed magnetic reso nance J Magn Reson A 1994 108 71 81 Rinard GA Quine RW Song R Eaton GR Eaton SS Absolute EPR spin echo and noise intensities J Magn Reson 1999 140 69 83 Ardenjaer Larsen JH Laursen I Leunbach I Ehnholm G Wistrand L G Petersson JS Golman K EPR and DNP properties of certain novel single electron contrast 54 55 56 57 58 59 60 61 agents intended for oximetric imaging J Magn Reson 1998 133 1 12 Quine RW Harbridge JR Eaton SS Eaton GR Design of a programmable timing unit Rev Sci Instrum 1999 70 4422 4432 Quine RW Programmable timing unit for generating multiple coherent timing signals US Patent 5 621 705 1997 Quine RW Programmable timing unit for generating multiple coherent timing signals US Patent 5 901 116 1999 Quine RW Rinard GA Eaton SS Eaton GR EPR resonator coupling monitor J Magn Reson 1992 99 571 575 Quine RW Eaton GR Setting the microwave phase in an EPR spectrometer J Magn Reson A 1996 119 268 270 Rinard GA Quine RW Eaton GR Eaton SS Adapting a Hall probe controller for current control of an air core magnet Concept
73. se of access to the sample from either the direction parallel with or perpendicular to the magnetic field In addi tion the 81 cm air core magnet was constructed with a nonmetallic support to minimize eddy currents dur ing rapid scans of the magnetic field Either the Bruker or DU system can control the 81 cm magnet while to date only the DU system controls the 40 cm magnet The choice of EPR signal detection scheme is also fundamental to the design of a spectrometer Tradi tionally CW EPR spectrometers operating in the 1 35 GHz range have used crystal detectors to rectify the microwave signal from the resonator High fre quency spectrometers commonly use bolometers Pulsed EPR spectrometers especially those designed for ESE or free induction decay FID detection have used a double balanced mixer DBM or quadrature mixer We have used DBM detection and quadrature mixer detection in our locally built spectrometers because the DBM retains phase information As de scribed in Refs 46 and 47 switching the phase of the Local Oscillator LO side of the DBM rapidly during pulse sequences makes it possible to perform rapid subtraction of signals in phase cycled pulse se quences A DBM also helps decrease sensitivity to phase noise from the source The quadrature mixer provides phase quadrature signals that can be com bined in computer postprocessing so that it is not essential to set the phase exactly prior to data acqui sition For som
74. shifter 64 is a four state phase shifter that allows shifting the phase of the power arm in 90 steps Pulse forming switch 65 forms the pulse High pass filter 66 atten uates video feedthrough induced by the switch At tenuators 53 and 54 provide 0 80 dB of attenuation by a combination of step and continuously variable controls Continuously variable attenuator 54 Wein schel model 910 20 11 has minimal phase shift with attenuation change Step attenuator 53 has consider able phase shift as a function of attenuation setting By operating in a range where only the continuously variable device has to be changed the phase can be kept reasonably constant Switch 97 provides a path to bring in a second RF frequency for use in electron electron double resonance experiments In the lower portion of the figure devices 67 56 and 57 are in the high power path after the external power amplifier Coupler 56 supplies signal to monitoring crystal 68 whose output is the incident power monitor signal that is selected on the front panel to go to the oscilloscope Crossed diode noise blanking circuit 67 Fig 16 is of local design and construction It uses small surface mount diodes type 1N4148W to provide a threshold of three diode levels in each direction The noise level of the power amplifier is well below this level so the diodes are cut off until a power pulse is applied During the power pulse the three diode drop at the zero crossing induces
75. so incorporates mag netic field gradient coils for three dimensional imag ing EPR signals are inherently weak and are especially weak at such a low RF frequency Consequently the spectrometer has many gain stages selectable for the type of measurement As shown in Figs 3 and 4 the signal from a CLR can be amplified immediately but 88 QUINE ET AL R12 R3 1 0K 4 02K an g er reais x 0150F 12V 15v Q2 NTE249 Gane RX OR 2N6057 22 910 i 15 HEAT SINK TEMP 2 Gur i H S 2 SENSOR c2 i 15V Lo 1UF 4 via Y LP347 TO FSDRIVE 9 a 2 4 FIGURE 27 3 i Lag R1 22 1K gt g 3 ANN c3 11 m 15V w INPUT DRIVE Q2 oo NTE250 DENOTES HIGH CURRENT PATH T7 OR 2N6050 HEAT SINK R10 R11 1 00K 1 00K R14 4 95K R16 R17 1 0K 9 09K ulD ULB LF347 LP347 13 va E 7 2 0 5 VOLT AMP TEMP t R15 H S 1 SENSOR tOn 4 99K RX z 110 1 412V 2 12V cx 3300F Pg 22UF 1sv fq GND 6 7 15V 8 Figure 28 Schematic diagram of linear sweep power amplifier GAIN ADJ R2 1 0K ari AW AXN I 20g CW FROM BRUKER ER 032 FIELD CONTROLLER 5 PIN DIN CONNECTOR RAPID SCAN INPUT 14 G V 12V TIME CONSTANT ADJ
76. ss corner at 43 mHz This low frequency response is necessary for stability given the high loop gain and the high Q of the resonant coil driver circuit Linear Ramp Field Sweep System A relatively fast compared to conventional slow scan data collection linear field sweep is often useful in setting up the spectrometer by observing the spectrum on an oscil loscope Our system provides a linear field sweep at 10 Hz for this purpose The ramp is produced by stage UIA Fig 17 The ramp is passed through stage UIB a 300 Hz low pass filter to limit the upper harmonic content Upper harmonic content must be suppressed so that it does not get through the 3 kHz high pass filter described above and into the modula tion feedback system The ramp is then scaled by stage U7C Fig 18 Stages U13 and U14 provide 12 bit digital amplitude control The digital value origi 80 QUINE ET AL c20 3KHz HIGH PASS CURRENT SENSOR RESONATOR PENETRATION c34 1 co 15 1UF 7 gaz Vt CALIBRATE X 02 OP37 oO 2 6 TP13 H gt T c35 4 co 15V TO FIGURE 26 Figure 19 nates in the PC from a user setting of desired field sweep amplitude The amplitude adjusted ramp is then sent out to the linear sweep power amplifier described below It is important to trim out the DC offset in st
77. the transfer function of the entire power amplifier is 2 A V The c79 cee op RPL 4 7K tage teso Y Eceo cel l csz 40 T 1ur T 1UF 7 1ur 19 vec e 10MHZ XTAUL ae 3 EREOR 1 e 7 al 9 ag 18 6 ca3 c84 ALE xrar pis e1 T1UF LUF PLASO Da3 P2303 P12 7 P12 eal P11 pit u35 pro plo g0c31 c83 577 MICROPROCESSOR 6 veo an cas aoe 27 7 p26 270256 28 T vec eR 33ut past 25 P25 27 aia i RST p245 P24 AALS ver sv P23 P23 24 Al2 sa p22 Bee 23a11 ont p7 p21 22 Pl js 2ta ostne 3 10 paol 21 P28 U39 c82 24 a5 ostt ps 74AC573 25 16 z jas oae pa 2 D7 9 12 LAB EA VDD ro7 en BaH Tane att o3 rosa D sjid 7a as o2 2 PSEN N posps4 257155 sq Tr er Sias oit p1 Poa 38 2 7 sD soHi capa al oono RXD PO3 aD ao a3 37 D2 4 17 LAB2 8 14 TxD P02 3D 3g a2 eno 38 Di 3 18 LABI 9 L rox 22 PTs Laso 10 gt ro o E Pe 1p Q AQ sate SeS EN OE INTI PSER d 2 cer INTO Mee oe aco e Ti WR 77 WRN 5V 5 C84 TO D RDN GND PSEN N v41 24 20 5V tice v22 44 ar0 vec to i 3 RER ACTIVITY sitt a b e 8 JP 2 k r204 RAM U36B LED i CY7C128A 5v
78. ts to the ring down exponential to estimate Q SpecJet is an eight bit digitizer with a 1 V dynamic range so a 10 mV signal is too small to be properly resolved and some ampli fication is needed Amplifier 97 provides a 20 dB 10X amplification to bring the signal up to the 100 mV level which fits the digitizer much better Ampli fier 97 is locally designed and constructed Plug in Configuration Modules Figures 6 and 7 show plug in configuration modules 17014 through 17017 These are small 9 4 cm X 4 6 cm microstrip circuit cards locally designed that plug into the bridge They reconfigure the resonator and circulator connections for the major operational modes of the bridge There are four different configuration mod ules CW LGR CW CLR pulse LGR and pulse CLR The modules are connectorized with SMB blind mate connectors and can be removed and in serted with the power on so long as the bridge is in STANDBY mode Microstrip coaxial switches on the pulse modules switch in CW power for resonator tuning purposes The configuration modules were de signed to eliminate additional switching paths that would have inserted additional losses and therefore noise into the signal paths In addition the use of plug in modules provides flexibility for ease of mod ifying these functions or adding new functions in the future AFC The AFC System consists of detector crystal 79 LGR mode or crystal 80 CLR mode followed by preamplifier 24
79. y 4 1 4 1N4148 1N414a I 1N4148 1N4148 1N4148 I mi D25 D40 D54 7 Hp 4 gt 4 1N4148 1N4148 1N4148 D27 D41 D55 H H 1N4148 1N4148 1N4148 D28 D42 D56 gt 1N4148 1N4148 1N4148 D29 D43 D57 li 4 1 p 1 1N4148 1N4148 1N4148 D30 D44 D58 gt bt 1n4148 1N4148 1N4148 Figure 16 Schematic diagram of crossed diode noise reduction circuit card eliminating noise We have not yet tested this feature although the printed circuit card incorporates the de sign The circuit card on which the noise blanking is implemented is 8 9 cm X 7 0 cm and has SMA connectors Power Amplifier Pending our acquisition of an appropriate power am plifier we have done pulsed EPR measurements with a temporary amplifier Dressler model 75A 250 It has fast rise time but no noise blanking and provides 25 W power at 250 MHz We plan to use a 400 500 W amplifier with noise blanking in the near future Microprocessor Controlled Interface Unit General The microprocessor controlled interface unit Figs 17 through 26 serves as an interface and control system between the PC and the major hard ware elements of the spectrometer such as the bridge modulation coils and modulation amplifier lock in amplifier magnet power supplies and oscilloscope display The unit contains one large printed circuit card of local design 40 cm X 35 cm on which all of the electronic components plus the input output con nectors and front
80. y A Kaplan N Callaghan PT Three dimensional pulsed ESR imaging J Magn Reson A 1996 121 201 205 Wokrina T Dormann E Kaplan N Conduction elec tron spin relaxation and diffusion in the radical cation salt diperylene hexafluorophosphate Phys Rev B 1996 54 10492 10501 Dormann E Sachs G St cklein W Bail B Schwoerer M Gaussmeter application of an organic conductor Appl Phys A 1983 30 227 231 Alexandrowicz G Tashma T Feintuch A Grayevsky A Dormann E Kaplan N Spatial mapping of mobility and density of the conduction electrons in FA PF Phys Rev Lett 2000 84 2973 2976 Feintuch A Alexandrowicz G Tashma T Boasson Y Grayevsky A Kaplan N Three dimensional pulsed ESR Fourier imaging J Magn Reson 2000 142 382 385 Alecci M Brivati JA Placidi G Sotgiu A A radiofre quency 220 MHz Fourier transform EPR spectrome ter J Magn Reson 1998 130 272 280 Alecci M Brivati JA Placidi G Test L Lurie DJ 29 30 31 32 33 34 35 36 37 38 39 40 41 42 43 44 45 Sotgiu A A Submicrosecond resonator and receiver system for pulsed magnetic resonance with large sam ples J Magn Reson 1998 132 162 166 Nishigawa H Fujii H Berliner LJ Helices and surface coils for low field in vivo ESR and EPR imaging applications J Magn Reson 1985 62 79 86 Berliner LJ Fujii H Magnetic resonance imaging of biological specimens by electron paramagnetic reso na
81. y phase noise is a less serious problem with a CLR than with a LGR One advantage of the CLR is that it isolates the EPR signal from the source noise by the amount of the isolation between the loops which can be at least 40 50 dB In one test at 250 MHz the noise was essentially constant with incident power when a CLR was used but increased sharply with increasing power with an otherwise similar LGR Future experiments may require a RF source with lower phase noise 68 QUINE ET AL CW CLR MODULE 17015 co cL 1000pf i a2 SOME COMPONENTS SHOWN MORE THAN ONCE FOR CLARITY t PULSE CLR MODULE 17016 FIGURE 5af s oe E i TO X 7 Ez COAKIAL SWITCH E F e Figure 7 CW Power Distribution CW power distribution in and connectors approximately 850 mW is available at the bridge is shown in Fig 5 The CW source is a the resonator A low power path is also available that Fluke generator external to the bridge supplying a bypasses the 1 W amplifier which supplies 50 mW 20 dBm power level The forward path to the res at the resonator This path is useful for experiments in onato

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